Metamaterial power amplifier systems

ABSTRACT

Power amplifying systems and modules and components therein are designed based on CRLH structures, providing high efficiency and linearity.

PRIORITY CLAIM AND RELATED APPLICATIONS

This patent application is a continuation of U.S. patent applicationSer. No. 12/708,437 entitled “Metamaterial Power Amplifier Systems,”filed on Feb. 18, 2010, which claims the benefit of the U.S. ProvisionalPatent Application Ser. No. 61/153,398 entitled “A Metamaterial PowerAmplifier System and Method for Generating Highly Efficient and LinearMulti-Band Power Amplifiers,” filed on Feb. 18, 2008. The disclosure ofthe above provisional application is incorporated herein by reference.This patent application is related to U.S. patent application Ser. No.11/741,674 entitled “Antennas, Devices and Systems based on MetamaterialStructures,” filed on Apr. 27, 2007; U.S. Pat. No. 7,592,952 entitled“Antennas Based on Metamaterial Structures,” issued on Sep. 22, 2009;and U.S. patent application Ser. No. 11/963,710 entitled “PowerCombiners and Dividers Based on Composite Right and Left HandedMetamaterial Structures,” filed on Dec. 21, 2007.

BACKGROUND

This document relates to power amplifier systems and components thereinbased on metamaterial structures.

The propagation of electromagnetic waves in most materials obeys theright-hand rule for the (E, H, β) vector fields, considering theelectrical field E, the magnetic field H, and the wave vector β (orpropagation constant). The phase velocity direction is the same as thedirection of the signal energy propagation (group velocity) and therefractive index is a positive number. Such materials are referred to asRight Handed (RH) materials. Most natural materials are RH materials.Artificial materials can also be RH materials.

A metamaterial (MTM) has an artificial structure. When designed with astructural average unit cell size much smaller than the wavelength ofthe electromagnetic energy guided by the metamaterial, the metamaterialcan behave like a homogeneous medium to the guided electromagneticenergy. Unlike RH materials, a metamaterial can exhibit a negativerefractive index, and the phase velocity direction is opposite to thedirection of the signal energy propagation, wherein the relativedirections of the (E, H, β) vector fields follow the left-hand rule.Metamaterials which have a negative index of refraction withsimultaneous negative permittivity ∈ and permeability μ are referred toas pure Left Handed (LH) metamaterials.

Many metamaterials are mixtures of LH metamaterials and RH materials andthus are Composite Right and Left Handed (CRLH) metamaterials. A CRLHmetamaterial can behave like an LH metamaterial at low frequencies andan RH material at high frequencies. Implementations and properties ofvarious CRLH metamaterials are described in, for example, Caloz andItoh, “Electromagnetic Metamaterials: Transmission Line Theory andMicrowave Applications,” John Wiley & Sons (2006). CRLH metamaterialsand their applications in antennas are described by Tatsuo Itoh in“Invited paper: Prospects for Metamaterials,” Electronics Letters, Vol.40, No. 16 (August 2004). CRLH metamaterials may be structured andengineered to exhibit electromagnetic properties tailored to specificapplications and may be used in applications where it may be difficult,impractical or infeasible to use other materials. In addition, CRLHmetamaterials may be used to develop new applications and to constructnew devices that may not be possible with RH materials.

In some applications, MTM and CRLH structures and components are basedon a technology which applies the concept of Left-handed (LH)structures. As used herein, the terms “metamaterial,” “MTM,” “CRLH,” and“CRLH MTM” refer to composite LH and RH structures engineered usingconventional dielectric and conductive materials to produce uniqueelectromagnetic properties, wherein such a composite unit cell is muchsmaller than the wavelength of the propagating electromagnetic waves.

Metamaterial technology, as used herein, includes technical means,methods, devices, inventions and engineering works which allow compactdevices composed of conductive and dielectric parts and are used toreceive and transmit electromagnetic waves. Using MTM technology,antennas and RF components may be made very compactly in comparison tocompeting methods and may be very closely spaced to each other or toother nearby components while at the same time minimizing undesirableinterference and electromagnetic coupling. Such antennas and RFcomponents further exhibit useful and unique electromagnetic behaviorthat results from one or more of a variety of structures to design,integrate, and optimize antennas and RF components inside wirelesscommunications devices.

CRLH structures are structures that behave as structures exhibitingsimultaneous negative permittivity (∈) and negative permeability (μ) ina frequency range and simultaneous positive ∈ and positive μ in anotherfrequency range. Transmission-line (TL) based CRLH structures arestructures that enable TL propagation and behave as structuresexhibiting simultaneous negative permittivity (∈) and negativepermeability (μ) in a frequency range and simultaneous positive ∈ andpositive μ in another frequency range. The CRLH based antennas and TLsmay be designed and implemented with and without conventional RF designstructures.

Antennas, RF components and other devices made of conventionalconductive and dielectric parts may be referred to as “MTM antennas,”“MTM components,” and so forth, when they are designed to behave as anMTM structure. MTM components may be easily fabricated usingconventional conductive and insulating materials and standardmanufacturing technologies including but not limited to: printing,etching, and subtracting conductive layers on substrates such as FR4,ceramics, LTCC, MMIC, flexible films, plastic or even paper.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an equivalent circuit of an MTM transmission line(TL) with at least three CRLH unit cells connected in series with aperiod p.

FIGS. 1A-1E illustrate various configurations of CRLH unit cells.

FIG. 1F illustrates a series RH TL expressed with an equivalent RH shuntcapacitance C_(R) and RH series inductance L_(R).

FIG. 1G illustrates a shunt RH TL expressed with an equivalent RH shuntcapacitance C′_(R) and RH shunt inductance L′_(R).

FIG. 2 illustrates a CRLH dispersion curve for a balanced CRLH unitcell, in comparison with an RH dispersion curve and an LH dispersioncurve.

FIG. 3 illustrates examples of an RH phase response, an LH phaseresponse and a CRLH phase response, indicated by dotted line,dashed-dotted line and solid line, respectively.

FIG. 4 illustrates an example of a conventional single-band bias circuitcoupled to an RF signal path for biasing a transistor in a poweramplifier (PA).

FIG. 5 plots simulation results of the impedance on the Smith Chart fortwo frequencies 2.4 GHz and 5.8 GHz for WiFi applications using theconventional single-band bias circuit illustrated in FIG. 4.

FIG. 6 illustrates an example of a dual-band bias circuit coupled to anRF signal path for biasing a transistor in a PA.

FIG. 7 plots measurement results of the impedance on the Smith Chart fortwo frequencies 2.4 GHz and 5.8 GHz using the dual-band bias circuitillustrated in FIG. 6.

FIG. 8 illustrates an example of a frequency selector having a shuntCRLH TL configuration.

FIG. 9 illustrates an example of a frequency selector having two CRLHTLs, one CRLH TL in shunt and the other in series.

FIG. 10 plots simulation results of the insertion loss for animplementation example of the frequency selector of FIG. 9.

FIG. 11 illustrates an example of an frequency selector having a seriesRH TL and a shunt CRLH TL.

FIG. 12 illustrates a layout of an implementation example of thefrequency selector of FIG. 11.

FIG. 13 plots simulation and measurement results of the return loss andinsertion loss of the implementation example of FIG. 12.

FIG. 14 illustrates an example of a frequency selector having a seriesCRLH TL and an open shunt RH TL.

FIG. 15 illustrates an example of a frequency selector having a seriesCRLH TL and a shorted shunt RH TL.

FIG. 16 illustrates an example of a frequency selector for passing asignal with one frequency and filtering out signals with two differentfrequencies.

FIG. 17 illustrates a layout of an implementation example of thefrequency selector of FIG. 16 with the port P3 being open.

FIG. 18 plots a phase response curve of the CRLH TL in theimplementation example of FIG. 17 in comparison with RH and LH responsecurves, indicated with the solid line, dotted line and dashed line,respectively.

FIG. 19 plots measurement and simulation results of return loss andinsertion loss for the implementation example of FIG. 17.

FIG. 20 illustrates an example of a frequency selector for passing asignal with the primary frequency and filtering out signals with the2^(nd) and 3^(rd) harmonics.

FIG. 21 illustrates an extended-CRLH (E-CRLH) unit cell.

FIG. 22 plots simulation results of insertion loss of a frequencyselector configured as in FIG. 16 but with an E-CRLH TL instead of aconventional-CRLH (C-CRLH) TL.

FIG. 23 illustrates an example of a multiband frequency selector havingseries elements X₁, X₂, . . . , X_(n) and shunt elements Y₁, Y₂, . . . ,Y_(n).

FIG. 24 illustrates an example of a multiband frequency selector havingtwo shunt elements coupled to each branch, as indicated by Y₁, Y₂, . . ., Y_(n) and Y_(1a), Y_(2a), . . . , Y_(na).

FIG. 25 illustrates a layout of an implementation example a diplexerhaving a frequency selector and a high-pass filter (HPF).

FIG. 26 plots measurement results of insertion loss P2-P1, insertionloss P3-P1 and isolation P3-P2 for the implementation example of FIG.25.

FIGS. 27A-27C illustrates three different ON/OFF configurations of twofrequency selectors connected in parallel.

FIG. 28 illustrates an example of an active frequency selector havingtwo CRLH TLs connected in shunt.

FIG. 29 illustrates an example of an active frequency selector havingtwo CRLH TLs connected in shunt and used as an active harmonic trap.

FIG. 30 illustrates a block diagram of a first power amplificationsystem for multiple frequency bands.

FIG. 31 illustrates a block diagram of a second power amplificationsystem for multiple frequency bands.

FIG. 32 illustrates an example of a dual-band PA based on the poweramplification system of FIG. 30 using active frequency selectors.

FIG. 33 illustrates an example of a dual-band PA based on the poweramplification system of FIG. 30 and having two input ports and twooutput ports.

FIG. 34 illustrates an example of a dual-band PA based on the poweramplification system of FIG. 30 and having two input ports and twooutput ports.

FIG. 35 illustrates I-V characteristics of a MESFET with bias points ofvarious class operations.

FIG. 36A-36C illustrate schematic diagrams of three differentconfigurations for removing harmonics in a PA.

FIG. 37 illustrates a layout of an implementation example of the class JMTM PA with the MTM harmonic trap of FIG. 36B.

FIG. 38 plots measurement results of Power Added Efficiency (PAE) andoutput power (Pout) as a function of input power (Pin) of theimplementation example of FIG. 37.

FIG. 39 illustrates a layout of an implementation example of the class-JMTM PA with the output matching impedance (OMN)-integrated MTM harmonictrap of FIG. 36C.

FIG. 40 plots measurement results of Pout versus Pin for theimplementation example of FIG. 39.

FIG. 41 plots measurement results of PAE versus Pout for theimplementation example of FIG. 39.

FIG. 42 illustrates measurement results of PAE versus Pin for theimplementation example of FIG. 39.

FIG. 43 plots simulation results of PAE versus frequency for amonolithic microwave integrated circuit (MMIC) implementation of theclass-J MTM PA of FIG. 39.

FIG. 44 illustrates a schematic diagram of a power amplifierconfiguration example using a CRLH TL with a varactor.

DETAILED DESCRIPTION

In modern communication systems, it is generally preferred that poweramplifiers (PAs) have high linearity and/or efficiency in order to meetvarious specifications and achieve a certain performance level. Highefficiency is important to prolong the battery lifetime of handsets sothat the handsets work for a longer period of time. High linearity isimportant to maintain the integrity of the signal with minimaldistortion.

MTM structures may be used to construct antennas, transmission lines andother RF components and devices, allowing for a wide range of technologyadvancements such as functionality enhancements, size reduction andperformance improvements. This document describes designs of PAs andcomponents used therein by using MTM structures to achieve both highefficiency and high linearity.

An MTM structure has one or more unit cells. The MTM-based componentsand devices may be designed based on these unit cells that may beimplemented by using distributed circuit elements, lumped circuitelements or a combination of both. FIG. 1 illustrates an equivalentcircuit 10 of an MTM transmission line (TL) with at least three CRLHunit cells connected in series with a period p. Denoting the length ofthe MTM TL by l and the number of CRLH unit cells by N, the relationshipof l=N×p holds in general. The equivalent circuit for each unit cell 12has an RH series inductance L_(R), an RH shunt capacitance C_(R), an LHseries capacitance C_(L) and an LH shunt inductance L_(L). The LH shuntinductance L_(L) and the LH series capacitance C_(L) may be structuredand connected to provide the LH properties to the unit cell 12, whilethe RH series inductance L_(R) and the RH shunt capacitance C_(R) may bestructured and connected to provide the RH properties to the unit cell12.

FIGS. 1A-1E illustrate various configurations of CRLH unit cells. Aseries RH block 100 represents an RH TL, such as a conventionalmicrostrip, which may be equivalently expressed with the RH shuntcapacitance C_(R) 102 and the RH series inductance L_(R) 104, asillustrated in FIG. 1F. When an RH TL is used in a shunt configurationas illustrated in FIG. 1G, the shunt RH TL 106 may be equivalentlyexpressed by an RH shunt capacitance C′_(R) 108 and an RH shuntinductance L′_(R) 110. Note that when the series RH TL 100 and the shuntRH TL 106 are combined with a similar TL size, the L_(R) value is moredominant than the L′_(R) value, and the C′_(R) value is more dominantthan C_(R) value. This indicates that the combination of a series andshunt RH TLs still provide RH properties with one overall RH shuntcapacitance and one overall RH series inductance. FIGS. 1D and 1Eillustrates examples of the original CRLH unit cell in FIG. 1 with theC_(R) and L_(R) replaced with the RH TL. “RH/2” in FIGS. 1A-1C refers tothe length of the RH TL being divided by 2. FIG. 1A illustrates asymmetric representation of the CRLH unit cell illustrated in FIG. 1with the C_(R) and L_(R) replaced with the RH TL, which is divided intotwo RH/2 for the symmetricity. Variations include a configuration asshown in FIG. 1A but with RH/2 and C_(L) interchanged; andconfigurations as shown in FIGS. 1A-1C but with RH/4 on one side and3RH/4 on the other side instead of RH/2 on both sides. Alternatively,other complementary fractions may be used to divide the RH transmissionline. The MTM structures may be implemented based on these CRLH unitcells by using distributed circuit elements, lumped circuit elements ora combination of both. Such MTM structures may be fabricated on variouscircuit platforms, including circuit boards such as a FR-4 PrintedCircuit Board (PCB) or a Flexible Printed Circuit (FPC) board. Examplesof other fabrication techniques include thin film fabricationtechniques, system on chip (SOC) techniques, low temperature co-firedceramic (LTCC) techniques, monolithic microwave integrated circuit(MMIC) techniques, and MEMS (Micro-Electro Mechanical System)techniques. Some examples and implementations of antenna structures,transmission lines and other RF components based on MTM structures aredescribed in the U.S. patent applications: Ser. No. 11/741,674 entitled“Antennas, Devices and Systems Based on Metamaterial Structures,” filedon Apr. 27, 2007; the U.S. Pat. No. 7,592,957 entitled “Antennas Basedon Metamaterial Structures,” issued on Sep. 22, 2009; and the U.S.patent application Ser. No. 11/963,710 entitled “Power Combiners andDividers Based on Composite Right and Left Handed MetamaterialStructures,” filed on Dec. 21, 2007.

A pure LH metamaterial follows the left-hand rule for the vector trio(E, H, β), and the phase velocity direction is opposite to the signalenergy propagation direction. Both the permittivity ∈ and permeability μof the LH material are simultaneously negative. A CRLH metamaterialexhibits left-handed and right-handed electromagnetic propertiesdepending on the regime or frequency of operation. In addition, the CRLHstructure may exhibit a non-zero group velocity when the wavevector β(or propagation constant) of a signal is zero. In an unbalanced casewhere C_(R)L_(L)≠C_(L)L_(R), there is a bandgap in which electromagneticwave propagation is inhibited. In a balanced case whereC_(R)L_(L)=C_(L)L_(R), the dispersion curve does not show discontinuityat the transition point of the propagation constant β(ω_(o))=0 betweenthe LH and RH regions, where the guided wavelength is infinite, i.e.,λ_(g)=2π/|β|→∝, while the group velocity is positive:

$\begin{matrix}{v_{g} = \left. \frac{\mathbb{d}\omega}{\mathbb{d}\beta} \middle| {}_{\beta = 0}{> 0.} \right.} & {{Eq}.\mspace{14mu}(1)}\end{matrix}$This state corresponds to the zeroth order mode in a TL implementation.

FIG. 2 illustrates a CRLH dispersion curve β for a balanced CRLH unitcell, where C_(R)L_(L)=C_(L)L_(R), in comparison to the RH dispersioncurve β_(R) and the LH dispersion curve β_(L). The CRLH dispersion curvefor this case may be approximated by:

$\begin{matrix}{\beta = {\frac{1}{p}{\left( {{\omega\sqrt{L_{R}C_{R}}} - \frac{1}{\omega\sqrt{L_{L}C_{L}}}} \right).}}} & {{Eq}.\mspace{14mu}(2)}\end{matrix}$In the unbalanced case where C_(R)L_(L)≠C_(L)L_(R), the dispersion curveβ may be expressed as:

$\begin{matrix}{{\beta = {\frac{1}{p}\left( {{s(\omega)}\sqrt{{\omega^{2}L_{R}C_{R}} + \frac{1}{\omega^{2}L_{L}C_{L}} - \left( {\frac{L_{R}}{L_{L}} + \frac{C_{R}}{C_{L}}} \right)}} \right)}},} & {{Eq}.\mspace{14mu}(3)}\end{matrix}$where

$\begin{matrix}{{s(\omega)} = \left\{ \begin{matrix}{- 1} & {{{if}\mspace{14mu}\omega} < {{\min\left( {\omega_{se},\omega_{sh}} \right)}\text{:}\mspace{14mu}{LH}\mspace{14mu}{range}}} \\{+ 1} & {{{if}\mspace{14mu}\omega} > {{\max\left( {\omega_{se},\omega_{sh}} \right)}\text{:}\mspace{14mu}{RH}\mspace{14mu}{{range}.}}}\end{matrix} \right.} & {{Eq}.\mspace{14mu}(4)}\end{matrix}$In the unbalanced case, there are two possible zero^(th) orderresonances, ω_(se) and ω_(sh), which can support an infinite wavelength(β=0, fundamental mode) and are expressed as:

$\begin{matrix}{\omega_{sh} = {{\frac{1}{\sqrt{C_{R}L_{L}}}\mspace{14mu}{and}\mspace{14mu}\omega_{se}} = {\frac{1}{\sqrt{C_{L}L_{R}}}.}}} & {{Eq}.\mspace{14mu}(5)}\end{matrix}$At ω_(se) and ω_(sh), the group velocity (v_(g)=dω/dβ) is zero and thephase velocity (v_(p)=ω/β) is infinite. When the CRLH unit cell isbalanced, these resonant frequencies coincide as shown in FIG. 2 and areexpressed as:ω_(se)=ω_(sh)=ω₀,  Eq. (6)where the positive group velocity (v_(g)=dω/dβ) as in Eq. (1) and theinfinite phase velocity (v_(p)=ω/β) may be obtained. In RH TLresonators, the resonance frequencies correspond to electrical lengthsof θ_(m)=β_(m)l=mπ (m=1, 2, 3, . . . ), where l is the length of the TL.The RH dispersion curve and the LH dispersion curve are represented byβ_(R) and β_(L), respectively, in FIG. 2. The CRLH dispersion curve isrepresented by β=β_(R)+β_(L) in FIG. 2, where β_(R) and β_(L) correspondto the first term and the second term, respectively, in Eq. (2). In CRLHTL resonators, the resonance frequencies correspond to electricallengths of θ_(m)=β_(m)l=mπ, where l is the length of the CRLH TL withthe relationship of l=N×p, and the parameter m=0, ±1, ±2, ±3, . . . ,±∝. Thus, a CRLH structure supports a spectrum of resonant frequencieswith a dispersion curve that extends to both negative and positive βregion, as indicated by the vertical lines intercepting the CRLHdispersion curve in FIG. 2.

For the balanced case having the dispersion curve expressed as in Eq.(2), the phase response may be expressed by:

$\begin{matrix}{{\phi_{CRLH} = {{\phi_{RH} + \phi_{LH}} = {{- \beta}\; 1}}},} & {{Eq}.\mspace{14mu}(7)} \\{{\phi_{RH} = {{- N}\; 2\;\pi\; f\sqrt{L_{R}C_{R}}}},} & {{Eq}.\mspace{14mu}(8)} \\{{\phi_{LH} = \frac{N}{2\pi\;\sqrt{L_{L}C_{L}}}},} & {{Eq}.\mspace{14mu}(9)}\end{matrix}$where l=N×p. Thus, the slope of the CRLH phase is given by:

$\begin{matrix}{\frac{\mathbb{d}\phi_{CRLH}}{\mathbb{d}f} = {{{- N}\; 2\pi\sqrt{L_{R}C_{R}}} - {\frac{N}{2\pi\; f^{2}\sqrt{L_{L}C_{L}}}.}}} & {{Eq}.\mspace{14mu}(10)}\end{matrix}$The characteristic impedance is given by:

$\begin{matrix}{Z_{o}^{CRLH} = {\sqrt{\frac{L_{R}}{C_{R}}} = {\sqrt{\frac{L_{L}}{C_{L}}}.}}} & {{Eq}.\mspace{14mu}(11)}\end{matrix}$Therefore, the equivalent circuit parameter values, C_(R), L_(L), C_(L)and L_(R) as well as the number of unit cells, N, may be selected andcontrolled, with constraints such as the impedance matching conditionsof Eq. (11), to create a desired phase response curve in a CRLHstructure. Furthermore, a non-zero frequency may be obtained at 0 degreein a CRLH structure unlike in an RH structure.

The following provides examples of determining the equivalent circuitparameters of a dual-band MTM structure. Similar techniques may be usedto determine the parameters with three or more bands. In a dual-band MTMdesign, signal frequencies f₁ and f₂ representing the two bands may befirst selected for two different phase values: φ₁ at f₁ and φ₂ at f₂.Using Eqs. (7)-(11), the values of parameters L_(R), C_(R), L_(L) andC_(L) can be obtained as:

$\begin{matrix}{{L_{R} = \frac{Z_{t}\left\lbrack {{\phi_{1}\left( \frac{\omega_{1}}{\omega_{2}} \right)} - \phi_{2}} \right\rbrack}{N\;{\omega_{2}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}}},{C_{R} = \frac{{\phi_{1}\left( \frac{\omega_{1}}{\omega_{2}} \right)} - \phi_{2}}{N\;\omega_{2}{Z_{t}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}}},{L_{L} = \frac{N\;{Z_{t}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}}{\;{\omega_{1}\left\lbrack {\phi_{1} - {\left( \frac{\omega_{1}}{\omega_{2}} \right)\phi_{2}}} \right\rbrack}}},{C_{L} = \frac{N\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}{\;{\omega_{1}{Z_{t}\left\lbrack {\phi_{1} - {\left( \frac{\omega_{1}}{\omega_{2}} \right)\phi_{2}}} \right\rbrack}}}},} & {{Eq}.\mspace{14mu}(12)}\end{matrix}$where Z_(t) is a given value for Z₀ ^(CRLH), e.g., 50Ω, representing thecharacteristic impedance of the underlying system. A CRLH TL has aphysical length of l with N unit cells, each having a period of p, wherel=N×p. The signal phase value is defined by φ=−βl. Therefore, we have:

$\begin{matrix}{{\beta = {- \frac{\phi}{l}}},} & {{Eq}.\mspace{14mu}(13)}\end{matrix}$which indicates:

$\begin{matrix}{{\beta_{i} = {- \frac{\phi_{i}}{\left( {N \cdot p} \right)}}},} & {{Eq}.\mspace{14mu}(14)}\end{matrix}$where i=1 or 2. It is thus possible to select two different phases φ₁and φ₂ at two different frequencies f₁ and f₂, respectively, as:

$\begin{matrix}\left\{ \begin{matrix}{\beta_{1} = {\frac{1}{p}\left( {{\omega_{1}\sqrt{L_{R}C_{R}}} - \frac{1}{\omega_{1}\sqrt{L_{L}C_{L}}}} \right)}} \\{\beta_{2} = {\frac{1}{p}{\left( {{\omega_{2}\sqrt{L_{R}C_{R}}} - \frac{1}{\omega_{2}\sqrt{L_{L}C_{L}}}} \right).}}}\end{matrix} \right. & {{Eq}.\mspace{14mu}(15)}\end{matrix}$Therefore, the five parameters L_(L), C_(R), L_(R), C_(L) and N asobtained above can determine the N resonant frequencies and phaseresponse curves, corresponding bandwidth, and input/output TL impedancevariations around these resonances. The above theory and derivations maybe found, for example, in Caloz and Itoh, “ElectromagneticMetamaterials: Transmission Line Theory and Microwave Applications,”John Wiley & Sons (2006).

FIG. 3 illustrates an example of an RH phase response, an LH phaseresponse and a CRLH phase response, indicated by dotted line,dashed-dotted line and solid line, respectively. The CRLH phase responsemay be expressed as a combination of the phase response of the RHcomponent and the phase response of the LH component as expressed inEqs. (7), (8) and (9). The CRLH TL phase response approaches to the LHphase response at low frequencies and approaches to the RH phaseresponse at high frequencies. Notably, the CRLH phase response curvecrosses the zero-phase axis with a frequency offset from zero andextends to both positive and negative infinity. Thus, the CRLH phaseresponse curve may be engineered to intercept a desired pair of phasesat an arbitrarily selected pair of non-zero frequencies. The values ofthe parameters L_(L), C_(R), L_(R), C_(L) and N may be selected andcontrolled to create a desired phase response. The number of unit cells,N, may be chosen to be a small number such as 1, 2 or 3 for achievingcircuit simplicity and component count reduction. With such a fixed Nvalue, the original four degree of freedom, i.e., L_(L), C_(R), L_(R),and C_(L), is reduced to two due to constraints such as the impedancematching conditions given in Eq. (11), in determining the parametervalues for the CRLH design. By way of example, FIG. 3 illustrates a CRLHdesign example in which the phase chosen at the first frequency f₁ is 0degree and the phase chosen at the second frequency f₂ is −360 degrees.

These CRLH properties may be utilized to provide antennas, directionalcouplers, matching networks, PAs, filters, power combiners andsplitters, and various other RF components and integrated systems forsingle-band as well as multi-band operations. Some implementationexamples of PAs and components or circuits used therein are describedbelow based on CRLH structures in comparison with conventional examples.

FIG. 4 illustrates an example 400 of a conventional single-band biascircuit coupled to an RF signal path for biasing a transistor in a PA.In RF communication systems, a bias circuit is generally designed todeliver DC power to the transistor while being transparent to the RFsignal so as to prevent the RF signal leakage into the bias circuit thatcan degrade the performance of the PA in which the bias circuit isincluded. The conventional single-band bias circuit of FIG. 4 includesthree RH TLs, 404, 408, and 412 connected radially at one end of eachTL, and coupled to the RF signal path 414 through a bias line 424. Thesingle band is represented by the operating frequency f₁, a bias voltageor current is supplied through the RH TL 404, and the RF signal path 414is indicated by the arrow through two circuit blocks 416 and 420. Thecircuit blocks 416 and 420 represent a transistor and other peripheralRF circuitry. The electrical length of each of the RH TLs 408 and 412may be chosen to be a quarter wavelength at f₁ to create a short circuitat the connected end (proximal end) and an open circuit at the other end(distal end). Equivalently, the phase φ₁ at f₁ of the RH TL 408 and thephase φ₂ at f₁ of the RH TL 412 may be chosen to be −90°×(2k+1), wherek=0, 1, 2, . . . . Note that the RH phase is negative except the DCpoint, as shown in Eq. (8) and FIG. 2. The locations of the short pointand the open point are indicated on the Smith Charts in FIG. 4. In thisexample, the distal end of the RH TL 408 is configured to be openinstead of being shorted to ground, a circuit or a component. By havinga quarter wavelength at f₁, the RH TL 408 acts as an impedancetransformer to change the impedance from an open at the distal end to ashort at the proximal end. Furthermore, by having a quarter wavelengthat f₁, the RH TL 412 acts as another impedance transformer to change theimpedance from a short at the proximal end to an open at the distal end,which is coupled to the transistor through a bias line 424. Therefore,the RF signal path 414 is not affected by the bias line 424 because itis open for the RF signal with f₁. This configuration allows for thesame operation at a single frequency or an odd multiple of a selectedfrequency based on the use of quarter-wavelength RH TLs. This is becausethe choice of the RH phases is restricted to be on the linear line infrequency, as shown in Eq. (8) and FIG. 2. Accordingly, the conventionaldesign based on RH TLs may not be suitable for a multiband application,which involves two or more arbitrary frequencies that may not be amultiple of each other.

In the above example, the shape of the quarter-wavelength RH TL ischosen to be a straight stub. In another conventional example, the shapeof the quarter-wavelength RH TL may be chosen to be radial, which maycontribute to increasing the bandwidth. However, such a radial RH TLgenerally occupies more real estate than a stub-shaped RH TL.

FIG. 5 plots simulation results of the impedance on the Smith Chart fortwo frequencies 2.4 GHz and 5.8 GHz for WiFi applications using theconventional single-band bias circuit illustrated in FIG. 4. The resultsindicate that it is unfeasible to have an open circuit in the bias line424 for two or more selected frequencies at the same time in theconventional single-band bias scheme. Specifically, the bias line mayform an open circuit at one of the frequencies of interest (e.g., 2.4GHz), but at the other frequency (e.g., 5.8 GHz) the bias line providesan impedance that may be at a different point on the Smith Chart asillustrated in FIG. 5. The difficulty of adjusting two impedancescorresponding to two arbitrary frequencies may be attributed to the useof conventional RH TLs, which have a substantially linear phase responseas shown by the dotted line in FIG. 3 and in Eq. (8).

Multiband bias circuits may be constructed based on MTM structures toovercome some of the deficiencies associated with the conventionalsingle-band bias circuit described as above. FIG. 6 illustrates anexample 600 of a dual-band bias circuit coupled to an RF signal path forbiasing a transistor in a PA. This dual-band bias circuit includes twoRH TLs 604 and 612 and one CRLH TL 608 connected radially at theproximal end of each TL, and coupled to the RF signal path 614 through abias line 624. The dual band is represented by the operating frequenciesf₁ and f₂, a bias voltage or current is supplied through the RH TL 604,and the RF signal path 614 with frequencies f₁ and f₂ is indicated bythe arrow through two circuit blocks 616 and 620. The circuit blocks 616and 620 represent a transistor and other peripheral circuitry. Thisconfiguration includes one CRLH TL 608 and one RH TL 612 for theimpedance transforming purpose, instead of two quarter-wavelength RH TLsas in the conventional single-band bias circuit in FIG. 4. Use of theCRLH TL 608 allows for an open bias line 624 at two different RFfrequencies f₁ and f₂ while permitting DC power to flow to thetransistor, as explained below with reference to the Smith charts in thefigure. The electrical length of the RH TL 612 may be chosen to have twoarbitrary phases φ₃=X° and φ₄=Y° at two frequencies f₁ and f₂,respectively, provided that they are in the linear relationship. Thatis, these two points correspond to a pair of points on the RH phaseresponse line such as the dotted line shown in FIG. 3. Due to thelinearity of the RH phase response, when one of the phases (X° or Y°) ischosen, the other (Y° or X°) is automatically fixed for fixed f₁ and f₂.In order to have an open circuit at the distal end of the RH TL 612 withthe distal end of the CRLH TL 608 being open, the impedances ZX and ZYcorresponding to the phases X° and Y° need to be compensated for by theCRLH TL 608. This is possible due to a greater degree of freedom indetermining the CRLH phase response than the RH phase response. Theelectrical length of the CRLH TL 608 may be chosen to have φ₁=360°−X° atf₁ and φ₂=360°−Y° at f₂, for example, so that the CRLH TL 608 acts as animpedance transformer between ZX/ZY and an open. These two pointscorrespond to a pair of points on the CRLH phase response line such asthe solid line shown in FIG. 3. The locations of the open point, the ZXpoint and the ZY point are indicated on the Smith Charts in FIG. 6. TheRF path 614 is not affected because the bias line 624 is open for the RFsignals with f₁ and f₂. In this example, the total electrical lengthfrom the open to the bias line 624 (i.e., from the open end of the CRLHTL 608 to the open end of the RH TL 612) is chosen to be 360°. Ingeneral, the total electrical length may be chosen to be 0°, 180°, 360°or a multiple of 180° to obtain the open-to-open impedancetransformation. That is, φ₁=k×180°−X° at f₁ and φ₂=k×180°−Y° at f₂,where k=0, ±1, ±2, . . . .

The dual-band MTM bias approach shown in FIG. 6 may be extended to amultiband MTM bias approach by choosing multiple phases for the RH TL612 corresponding to multiple frequencies on the RH phase response linesuch as the dotted line in FIG. 3, and choosing the CRLH phases tocompensate for the respective RH phases. Furthermore, this configurationmay be extended to the case where the distal end of the CRLH TL 608 isshorted instead of open. In this case, the phases of the CRLH TL 608 maybe chosen so that the CRLH TL 608 acts as an impedance transformer totransform the short to ZX/ZY. In this example, the total electricallength from the short to the bias line 624 (i.e., from the shorted endof the CRLH TL 608 to the open end of the RH TL 612) may be chosen to beφ₁=k×90°−X° at f₁ and φ₂=k×90°−Y° at f₂, where k=±1, ±2, . . . . Thesimilar design approach based on the CRLH structure may be employed forthree or more band biasing to provide a multiband MTM bias circuit withthe shorted CRLH TL as well.

FIG. 7 plots measurement results of the impedance on the Smith Chart fortwo frequencies 2.4 GHz and 5.8 GHz using the dual-band bias circuitillustrated in FIG. 6. In this example, one CRLH unit cell asillustrated in FIG. 1 is used to construct the CRLH TL 608, with theequivalent circuit parameter values of about C_(L)=0.61 pF, L_(L)=5.25nH, C_(R)=1.96 pF and L_(R)=16.9 nH, which provide the adequate CRLHphase response for this dual-band application. The results indicate thatan open circuit may be obtained at the bias line for both f₁=2.4 GHz andf₂=5.8 GHz due to the design and phase flexibility by use of the CRLHTL.

In another example, a dual-band bias circuit based on a CRLH TL may beconfigured to have an open bias line for one frequency f₁ and a shortedbias line at a different frequency f₂. Alternatively, the secondfrequency f₂ may be selected to be a second or higher harmonic.

In the example 600 of the dual-band bias circuit design described above,the RH TL 612 and CRLH TL 608 are configured to block a single RF signalor double RF signals at the bias line. Instead of blocking, signals withcertain frequencies may be selected to pass through a desired path. Sucha frequency selector may be constructed using a CRLH structure, in whicha signal with one frequency f₁ is allowed to transmit while anothersignal with a different frequency f₂ is blocked or filtered out. Thefrequency selector may thus provide a filtering function, as typicallyexhibited in a notch filter, for example, and may be designed as abuilding block for a PA or other communication systems. Some examples offrequency selectors based on CRLH structures are given below.

FIG. 8 illustrates an example of a frequency selector 800 having a shuntCRLH TL configuration. A port P1 is an input port where RF signals areinputted, and a port P2 is an output port where the RF signals areoutputted. One end (distal end) of a CRLH TL 804 is coupled to a portP3, which may be shorted or open. The other end (proximal end) of theCRLH TL 804 is coupled to the signal path P1-P2 through a shunt line808. The shunt line 808 may be controlled to be open for the signal withf₁ and shorted for the signal with f₂ by configuring the CRLH TL 804 tohave adequate phases for the signals with f₁ and f₂ based on the CRLHphase response. In this example, the CRLH TL 804 is utilized as animpedance transformer that transforms the impedance Z3 (e.g., open orshort) at the port P3 to an open for f₁ and to a short for f₂ at theshunt line 808. As a result, the signal with f₁ is allowed to transmitthrough the P1-P2 path, while the signal with f₂ is filtered out orremoved through the shunt line 808 and the CRLH TL 804, being blockedfrom reaching the output port P2. Theoretically, the impedance Z3 isinfinite when the port P3 is open, and is zero when the port P3 isshorted. However, in actuality and in this document, the term “an open”is used to indicate a high impedance achievable for the implementationand application and the term “a short” is used to indicate a lowimpedance achievable for the implementation and application.

The electrical length of the CRLH TL 804 may be engineered to correspondto a certain phase at a certain frequency as illustrated with the CRLHphase response curve in FIG. 3. In order to have the shunt line 808 openat f₁ and the shunt line 808 shorted at f₂ when the port P3 is open, awide variety of phase combinations may be used. Using the CRLH TL, thephase φ₁ at f₁ may configured to be:φ₁=0°±(k×180°),  Eq. (16)where k=0, 1, 2, . . . , to have an open circuit at the shunt line 808.Similarly, the phase φ₂ at f₂ may be configured to be:φ₂=90°±(k×180°),  Eq. (17)where k=0, 1, 2, . . . , to have a shorted circuit at the shunt line808. Alternatively, when the port P3 is shorted, a wide variety of phasecombinations may also be used in order to have an open shunt line 808 atf₁ and a shorted shunt line 808 at f₂. For the case of having the shuntCRLH TL 804 shorted to ground, a circuit or a component, the phase φ₁ atf₁ may be configured to be:φ₁=90°±(k×180°),  Eq. (18)where k=0, 1, 2, . . . , to have an open circuit at the shunt line 808.Similarly, the phase φ₂ at f₂ may be configured to be:φ₂=0°±(k×180°),  Eq. (19)where k=0, 1, 2, . . . , to have a short circuit at the shunt line 808.

Based on the frequency selecting scheme using the shunt CRLH TL 804 asexplained above, various frequency selectors may be constructed usingcombinations of CRLH and RH TLs. For example, a frequency selector mayinclude a combination of two CRLH TLs, while another example may includea CRLH TL and an RH TL, to perform the frequency selecting function.FIG. 9 illustrates an example of a frequency selector 900 having twoCRLH TLs, one CRLH TL 904 in shunt and the other 912 in series with thesignal path P1-P2. Similar to the configuration illustrated in FIG. 8,one end (distal end) of the shunt CRLH TL 904 is coupled to the port P3,which may be shorted or open. The other end (proximal end) of the shuntCRLH TL 904 is coupled to the signal path P1-P2 through a shunt line908. The shunt line 908 may be controlled to be open for the signal withf₁ and shorted for the signal with f₂ by configuring the shunt CRLH TL904 to have adequate phases φ₁ and φ₂ for the signals with f₁ and f₂,respectively. The phases φ₁ and φ₂ of the shunt CRLH TL 904 may bechosen from the values in Eqs. (16) and (17), respectively, when the P3is open and in Eqs. (18) and (19) when the P3 is shorted. One end of theseries CRLH TL 912 is coupled to the input port P1 and the other end iscoupled to the shunt line 908. The phases φ₃ and φ₄ of the series CRLHTL 912 may be selected as below when the port P3 is either open orshorted. For the signal with f₁, the shunt CRLH TL 904 is decoupled fromthe signal path P1-P2; thus, the series CRLH TL 912 may be structured tohave a matched impedance between the input and output impedances, e.g.,50Ω, with the phase φ₃ at f₁ being arbitrary. For the signal with f₂,the shunt line 908 is shorted, and the input port P1 is preferablydesigned to be open to have maximum signal reflection. Thus, the φ₄ atf₂ of the series CRLH TL 912 may be chosen to be 90°±(k×180°), wherek=0, 1, 2, . . . , to transform the impedance from a short to an open.Alternatively, to design the input port P1 to be shorted, the φ₄ at f₂of the series CRLH TL 912 may be chosen to be 0°±(k×180°), where k=0, 1,2, . . . , to transform or keep the impedance from a short to a short.Furthermore, the CRLH TL 904 may be designed differently so that thesephases φ₃ and φ₄ have other values suited for underlying applications toadjust the signal transmission and reflection. Using a series TL and ashunt TL, as in the above example, provides the flexibility to adjustthe phases φ₃ and φ₄ associated with the series TL while the phases φ₁and φ₂ associated with the shunt TL are kept to provide an open or ashort to block or transmit signals with certain frequencies.

In one implementation example, the frequency selector 900 of FIG. 9 maybe constructed to have the series CRLH TL 912 providing a phase φ₃=0° ata frequency f₁=2.4 GHz and a phase φ₄=−90° at a frequency f₂=5.8 GHz andthe shunt CRLH TL 904 providing a phase φ₁=0° at a frequency f₁=2.4 GHzand a phase φ₂=−90° at a frequency f₂=5.8 GHz with the port P3 open.These phase responses may be obtained by designing the series CRLH TL912 with two CRLH unit cells and the shunt CRLH TL 904 with one CRLHunit cell, each unit cell having equivalent circuit parameter values ofabout C_(L)=1.7 pF, L_(L)=4.25 nH, L_(R)=2.6 nH and C_(R)=1 pF.

FIG. 10 plots simulation results of insertion loss for an implementationexample of the frequency selector of 900 of FIG. 9 with the aboveequivalent circuit parameter values based on an FR4 substrate. The plotindicates that the insertion loss at 2.4 GHz, indicated by the point m3,is close to zero, and the insertion loss at 5.8 GHz, indicated by thepoint m4, is close to −30 dB. Therefore, these results indicate thatsignals around 2.4 GHz pass through the frequency selector 900 whilesignals around 5.8 GHz are blocked.

FIG. 11 illustrates another example of a frequency selector 1100. Thisfrequency selector has a series RH TL 1112 and a shunt CRLH TL 1104. Ingeneral, use of an RH TL allows for control of the phase at onefrequency; the phase at a second frequency may not be chosen arbitrarilybecause of the linearity of the RH phase response. As described earlier,to pass the signal with frequency f₁ from the port P1 to P2 and blockthe signal with frequency f₂ by filtering out the signal through theport P3, the phases φ₁ and φ₂ of the shunt CRLH TL 1104 may be chosen asexpressed by Eqs. (16) and (17) when the port P3 is open, and asexpressed by Eqs. (18) and (19) when the port P3 is shorted. The phasesφ₃ and φ₄ of the series RH TL 1112 may be selected as below when theport P3 is either open or shorted. For the signal with f₁, the shuntCRLH TL 1104 is decoupled from the signal path P1-P2; thus, the seriesRH TL 1112 may be structured to have a matched impedance between theinput and output impedances, e.g., 50Ω, with the phase φ₃ at f₁ beingarbitrary. For the signal with f₂, the shunt line 1108 is shorted, andthe input port P1 is preferably designed to be open to have maximumsignal reflection. Thus, the φ₄ at f₂ of the series RH TL 1112 may bechosen to be −90°−(k×180°), where k=0, 1, 2, . . . , to transform theimpedance from a short to an open. In general, the RH phase values arerestricted to be negative for non-DC frequencies. In this example, thephase φ₃ may be arbitrary as long as it is on the linear RH phaseresponse determined by the φ₄ value at f₂. Alternatively, to design theinput port P1 to be shorted, the φ₄ at f₂ of the series RH TL 1112 maybe chosen to be 0°−(k×180°), where k=1, 2, . . . , to transform or keepthe impedance from a short to a short. Furthermore, the RH TL 1112 maybe designed differently so that these phases φ₃ and φ₄ have other valuessuited for underlying applications to adjust the signal transmission andreflection.

In one implementation example based on the configuration of thefrequency selector 1100, the shunt CRLH TL 1104 for the case of the openport P3 may be constructed with φ₁=0° at f₁=2.4 GHz and φ₂=−90° atf₂=5.8 GHz. For the series RH TL 1112, the phase φ₄ at f₂=5.8 GHz ischosen to be −90° and the phase φ₃ at f₁=2.4 GHz may take on anarbitrary value as long as the two points are on the same RH phaseresponse line. Using a FR4 substrate with a thickness of about 31 milsand a dielectric constant of about 4.4, the phase φ₃ at 2.4 GHz may bechosen to be about −37° when the phase φ₄ at 5.8 GHz is −90°.

FIG. 12 illustrates a layout 1200 of the above implementation example ofthe frequency selector 1100 of FIG. 11. In this layout, the white partrepresents conductive patches and lines printed on the FR4 substrate,while the black part represents dielectric gaps formed between thoseconductive patches and lines. The shunt CRLH TL 1204 is designed to haveone CRLH unit cell with C_(L)=2 pF, L_(L)=5 nH, C_(R)=1 pF and L_(R)=2.6pF. In this example, two lumped capacitors, each having 2C_(L), areconnected in series to provide one C_(L) as in FIG. 1A. Alternatively, asingle capacitor may be used to provide a capacitance value of C_(L) inplace of two series capacitors. Furthermore, this example has twomicrostrips connected to form an L-shaped RH TL (L_(L)) 1216 to provideL_(L). Alternatively, a single microstrip having the same totalelectrical length may be used. In addition, a lumped inductor may beused to realize L_(L) instead of the printed microstrip. The RH portionof the shunt CRLH TL 1204, i.e., C_(R) and L_(R), is implemented using amicrostrip to provide an RH TL (C_(R), L_(R)) 1208 as shown in FIG. 1FThe RH TL 1212 is also implemented using a microstrip in this example.

FIG. 13 plots simulation and measurement results of return loss andinsertion loss of the above implementation example 1200 FIG. 12. Forboth simulation and measurement, the insertion loss has a dip near 5 GHzand is close to zero around 2.4 GHz. This indicates that the signalsaround 2.4 GHz are transmitted through the frequency selector with smallsignal loss. In contrast, the signals around 5 GHz are blocked fromtransmitting through the same frequency selector. The return loss forboth simulation and measurement has a dip near 2.4 GHz and close to zeronear 5 GHz, indicating that a good resonance is obtained around 2.4 GHzwith minimal signal reflection at the input port P1 and nearly fullsignal reflection is obtained around 5 GHz providing a near-open inputport P1.

FIG. 14 illustrates another example of a frequency selector 1400. Thisfrequency selector 1400 includes a series CRLH TL 1412 and a shunt RH TL1404 that is open at P3. In this example, the signal with f₂ is allowedto transmit from the port P1 to the port P2 while the signaltransmission with f₁ is blocked by filtering out through the port P3.The phase φ₁ at f₁ of the shunt RH TL 1404 may be chosen fromφ₁=−90°−(k×180°), where k=0, 1, 2, . . . , to transform the impedancefrom an open to a short so as to have a shorted shunt line 1408 for f₁.Since the RH phase is linear in frequency, the selection of the phasesφ₁ at f₁ and φ₂ at f₂ is limited. One simple selection is to have f₂ tobe a harmonic of f₁. By choosing φ₁=−90°, the phase φ₂ at f₂ of theshunt RH TL 1404 may be chosen from φ₂=2n×(−90°) at f₂=(n+1)×f₁, wheren=1, 2, 3, . . . , to have an open shunt line 1408 for the signal withf₂ to pass from P1 to P2. The phases φ₃ and φ₄ of the series CRLH TL1412 may be determined as follows. For the signal with f₁, the shuntline 1408 is shorted, and the input port P1 is preferably designed to beopen to have maximum signal reflection. Thus, the φ₃ at f₁ of the seriesCRLH TL 1412 may be chosen from 90°±(k×180°), where k=0, 1, 2, . . . ,to transform the impedance from a short to an open. For the signal withf₂, which is n×f₁ with n being a positive even number in this case, theshunt RH TL 1404 is decoupled from the signal path P1-P2; thus, theseries CRLH TL 1412 may be structured to have a matched impedancebetween the input and output impedances, e.g., 50Ω, with the phase φ₄ atf₂ being arbitrary. Alternatively, to design the input port P1 to beshorted, the φ₃ at f₁ of the series CRLH TL 1412 may be chosen to be0°±(k×180°), where k=1, 2, . . . , to transform or keep the impedancefrom a short to a short. Furthermore, the CRLH TL 1412 may be designeddifferently so that these phases φ₃ and φ₄ have other values suited forunderlying applications.

FIG. 15 illustrates another example of a frequency selector 1500. Thisfrequency selector 1500 includes a series CRLH TL 1512 and a shunt RH TL1504 that is shorted to ground. The phase φ₁ at f₁ of the shunt RH TL1504 may be chosen from φ₁=−(k×180°), where k=1, 2, . . . , to transformor keep the impedance from a short to a short so as to have a shortedshunt line 1508 to filter out the signal with f₁ through the port P3.Since the RH phase is linear in frequency, the selection of the phasesφ₁ at f₁ and φ₂ at f₂ is limited. One simple selection is to have f₁ tobe a harmonic of f₂. By choosing φ₁=−180°, the phase φ₂ at f₂ of theshunt RH TL 1504 may be chosen from φ₂=(2n−1)×(−90° at f₂=(n+1)×f₁,where n=1, 2, 3, . . . , to have an open shunt line 1508 for the signalwith f₂ to pass from P1 to P2. The phases φ₃ and φ₄ of the series CRLHTL 1512 may be determined as follows. For the signal with f₁, the shuntline 1508 is shorted, and the input port P1 is preferably designed to beopen to have maximum signal reflection. Thus, the φ₃ at f₁ of the seriesCRLH TL 1512 may be chosen from 90°±(k×180°), where k=0, 1, 2, . . . totransform the impedance from a short to an open. For the signal with f₂,which is (n+1)×f₁ in this case, the shunt RH TL 1504 is decoupled fromthe signal path P1-P2; thus, the series CRLH TL 1512 may be structuredto have a matched impedance between the input and output impedances,e.g., 50Ω, with the phase φ₄ at f₂ being arbitrary. Alternatively, todesign the input port P1 to be shorted, the φ₃ at f₁ of the series CRLHTL 1512 may be chosen to be 0°±(k×180°), where k=1, 2, . . . , totransform or keep the impedance from a short to a short. Furthermore,the CRLH TL 1512 may be designed differently so that these phases φ₃ andφ₄ have other values suited for underlying applications.

Referring back to FIG. 8, the frequency selector 800 has a shunt CRLH TL804 which is designed to provide phases at two different frequencies f₁and f₂ such that a signal with one frequency f₁ is allowed to passthrough the path P1-P2 while another signal with a different frequencyf₂ is blocked. It is possible to design the shunt CRLH TL to providephases at three different frequencies such that a signal with onefrequency is allowed to pass through the path P1-P2 while the signalswith the other two frequencies are blocked. FIG. 16 illustrates anexample of a frequency selector 1600 for passing a signal with frequencyf₂ through the path P1-P2 and filtering out a signal with frequency f₁and a signal with frequency f₃. One end (distal end) of a CRLH TL 1604is coupled to a port P3, which may be shorted or open. The other end(proximal end) of the CRLH TL 1604 is coupled to the signal path P1-P2through a shunt line 1608. The shunt line 1608 may be controlled to beopen for the signal with f₂ and shorted for the signals with f₁ and f₃by configuring the CRLH TL 1604 to have adequate phases for the signalswith f₁, f₂ and f₃ based on the CRLH phase response. In this example,the CRLH TL 1604 is utilized as an impedance transformer that transformsthe impedance Z3 (e.g., open or short) at the port P3 to an open for f₂and to a short for f₁ and f₃ at the shunt line 1608. As a result, thesignal with f₂ is allowed to transmit through the P1-P2 path, while thesignal with f₁ and the signal with f₃ are filtered out or removedthrough the shunt line 1608 and the CRLH TL 1604, being blocked fromreaching the output port P2. Theoretically, the impedance Z3 is infinitewhen the port P3 is open, and is zero when the port P3 is shorted.However, in actuality and in this document, the term “an open” is usedto indicate a high impedance achievable for the implementation andapplication and the term “a short” is used to indicate a low impedanceachievable for the implementation and application.

The electrical length of the CRLH TL 1604 may be engineered tocorrespond to certain phases at certain frequencies as shown in the CRLHphase response curve in FIG. 3. In order to have the shunt line 1608open at f₂ when the port P3 is open, the phase φ₂ at f₂ may beconfigured to be: φ₂=0°±(k×180°) where k=0, 1, 2, . . . , as in Eq.(16). Similarly, in order to have the shunt line 1608 shorted at f₁ andf₃, the phase φ₁ at f₁ and the phase φ₃ at f₃ may be configured to be:φ₁, φ₃=90°±(k×180°) where k=0, 1, 2, . . . , as in Eq. (17).Alternatively, when the port P3 is shorted, the phase φ₂ at f₂ may beconfigured to be: φ₂=90°±(k×180°) where k=0, 1, 2, . . . , as in Eq.(18) to have an open circuit at the shunt line 1608. Similarly, thephase φ₁ at f₁ and the phase φ₃ at f₃ may be configured to be: φ₁,φ₃=0°±(k×180°), where k=0, 1, 2, . . . , as in Eq. (19) to have ashorted circuit at the shunt line 1608.

In some WiFi applications such as for 802.11b, g, and n, it is desirableto remove the Digital Enhanced Cordless Telecommunication (DECT) bandranging 1880-1900 MHz (referred to as the 1.9 GHz band hereinafter) aswell as the 5 GHz band in one branch path, and pass the 2.4 GHz bandranging 2.4-2.48 GHz in the other branch path. A conventional diplexertypically uses one low-pass filter (LPF) in one branch and one high-passfilter (HPF) in the other branch. In the present case, the band thatneeds to pass is the 2.4 GHz band and the bands that need to be blockedare the 1.9 GHz band and the 5 GHz band, giving the order ofblock-pass-block. Thus, the conventional diplexer may not be suited forhandling such a complex filtering due to the simple combination of a HPFand a LPF, which generally gives the order of pass-block-pass.

The frequency selector 1600 of FIG. 16 provides a double-notch filterfunction, and thus may be configured to remove the 1.9 band and the 5GHz band and pass the 2.4 GHz band. This may be achieved with a compactdesign printed on an FR4 substrate based on CRLH structures, therebyleading to size reduction and hence power loss reduction.

FIG. 17 illustrates a layout of an implementation example 1700 of thefrequency selector 1600 of FIG. 16 with the port P3 being open. Thislayout is similar to the frequency selector 1200, having a shunt CRLH TL1704 with one symmetric CRLH unit cell and a series RH TL 1 1708. Thisimplementation example 1700 is designed to remove the 1.9 GHz band andthe 5 GHz band, and pass the 2.4 GHz band. In this example, two lumpedcapacitors, each having 2C_(L), are connected in series to provide oneC_(L) as in FIG. 1A. Furthermore, this CRLH unit cell has twomicrostrips connected to form an L-shaped RH TL (L_(L)) 1712 to provideL_(L). The RH portion of the shunt CRLH TL 1704, i.e., C_(R) and L_(R),is implemented using a microstrip to provide an RH TL (C_(R), L_(R))1716 as shown in FIG. 1F. The series RH TL 1708 is also implementedusing a microstrip in this example. The CRLH parameter values are chosento be: L_(L)=4 nH with a length of 7 mm, C_(L)=1 pF, L_(R)=2.66 nH, andC_(R)=1 pF. The substrate used is FR4 with a thickness of 10 mils.

FIG. 18 plots the phase response curve of the CRLH TL 1704 in FIG. 17having the above parameter values in comparison with the RH and LHresponse curves, indicated with the solid line, dotted line and dashedline, respectively. In this implementation example 1700, the shunt CRLHTL 1704 with the open port P3 is structured with the above parametervalues to provide approximately φ₁=90° at f₁=1.9 GHz, φ₂=0° at f₂=2.4GHz, and φ₃=−90° at f₃=5 GHz, as indicated by the three points on theCRLH phase response curve in FIG. 18.

FIG. 19 plots measurement and simulation results of the return loss andinsertion loss for the above implementation example 1700 of thefrequency selector 1600. The plots indicate that this frequency selectorfilters out the signals around 1.9 GHz and 5 GHz while passing thesignals around 2.4 GHz, exhibiting the double-notch filtercharacteristics.

Referring back to FIG. 11, a series RH TL 1112 is used in this frequencyselector 1100 for controlling the RH phases for a certain application,in which an open circuit at the port P1 is preferred to obtain maximumsignal reflection. The similar configuration may be extended tocontrolling three phases at three different frequencies by incorporatingthe CRLH TL design as in FIG. 16 with a series RH TL such as the seriesRH TL 1112 in FIG. 11. FIG. 20 illustrates an example of a frequencyselector 2000 for passing a signal with the primary frequency f andfiltering out signals with the 2^(nd) and 3^(rd) harmonics 2f and 3f.The frequency selector 2000 has a shunt CRLH TL 2004 and a series RH TL2012, providing capability as a harmonic trap. One end (distal end) ofthe CRLH TL 2004 is coupled to the port P3, which may be shorted oropen. The other end (proximal end) of the CRLH TL 2004 is coupled to thesignal path P1-P2 through a shunt line 2008. The shunt line 2008 may becontrolled to be open for the signal with the primary frequency f andshorted for the signals with the harmonics 2f and 3f by configuring theCRLH TL 1604 to have adequate phases for the signals with f, 2f and 3fbased on the CRLH phase response. In order to have the shunt line 2008open at f when the port P3 is open, the phase φ₁ at f may be configuredto be: φ₁=0°±(k×180°) where k=0, 1, 2, . . . , as in Eq. (16).Similarly, in order to have the shunt line 2008 shorted at 2f and 3f,the phase φ₂ at 2f and the phase φ₃ at 3f may be configured to be: φ₂,φ₃=90°±(k×180°) where k=0, 1, 2, . . . , as in Eq. (17). Alternatively,when the port P3 is shorted, the phase φ₁ at f may be configured to be:φ₁=90°±(k×180°) where k=0, 1, 2, . . . , as in Eq. (18) to have an opencircuit at the shunt line 2008. Similarly, the phase φ₂ at 2f and thephase φ₃ at 3f may be configured to be: φ₂, φ₃=0°±(k×180°), where k=0,1, 2, . . . , as in Eq. (19) to have a short circuit at the shunt line2008. The phases φ₄ at f, φ₅ at 2f and φ₆ at 3f of the series RH TL 2012may be selected as below when the port P3 is either open or shorted. Forthe signal with f, the shunt CRLH TL 2004 is decoupled from the signalpath P1-P2; thus, the series RH TL 2012 may be structured to havematching impedance between the input and output impedances, e.g., 50Ω,with the phase φ₄ at f being arbitrary. For the signal with 2f and 3f,the shunt line 2008 is shorted, and the input port P1 is preferred to beopen for odd harmonics and shorted for even harmonics in certainapplications. Thus, the phase φ₅ at 2f of the series RH TL 2012 may bechosen to be 0°−(k×180°), where k=1, 2, . . . , to transform or keep theimpedance from a short to a short; and φ₆ at 3f of the series RH TL 2012may be chosen to be −90°−(k×180°), where k=0, 1, 2, . . . , to transformthe impedance from a short to an open. Note that RH phase values arerestricted to be negative and linear in frequency. Thus, for example,these three phases may be chosen to be φ₄=−90° at f, φ₅=−180° at 2f andφ₆=−270° at 3f based on the RH phase response.

As described in the design examples thus far, the equivalent circuitparameters are manipulated to fit the resultant CRLH phase responsecurve to two or three points representing desired two or three pairs ofa phase and a frequency. The passing and blocking of the signals in thetwo or three frequency bands are thus engineered by the chosen phases.The possible number of fitting points on the CRLH phase response curvedepends on the degree of freedom in the design parameters. Generally,the design flexibility and hence the number of fitting points increasesas the number of parameters increases. Use of one CRLH unit cell, forexample, gives four parameters C_(L), L_(L), C_(R), and L_(R), but theimpedance matching conditions such as in Eq. (11) and other conditionsmay reduce the number of adjustable parameters. As explained below, useof a different type of CRLH unit cells having more parameters is onepossible solution to increase the design flexibility and hence thenumber of fitting points on the CRLH phase response curve. As a result,a filter device that is capable of passing and blocking a multiplenumber of bands may be designed and implemented.

FIG. 21 illustrates an extended-CRLH (E-CRLH) unit cell 2100. Theoriginal CRLH unit cell 12 such as illustrated in FIG. 1 is termed aconventional-CRLH (C-CRLH) unit cell for comparison. An E-CRLH unit cell2100 may be defined as a combination of a C-CRLH unit cell and adual-CRLH (D-CRLH) cell with a period p. The C-CRLH series part 2104 hasan RH series inductor L^(c) _(R) and an LH series capacitor C^(c) _(L)that are coupled in series; the C-CRLH shunt part 2108 has an RH shuntcapacitor C^(c) _(R) and an LH shunt inductor L^(c) _(L) that arecoupled in parallel; the D-CRLH series part 2112 has an RH seriesinductor L^(d) _(R) and an LH series capacitor C^(d) _(L) that arecoupled in parallel; and the D-CRLH shunt part 2116 has an RH shuntcapacitor C^(d) _(R) and an LH shunt inductor L^(d) _(L) that arecoupled in series. Thus one E-CRLH unit cell 2100 provides 8 equivalentcircuit parameters, L^(c) _(R), C^(c) _(L), C^(c) _(R), L^(c) _(L),L^(d) _(R), C^(d) _(L), C^(d) _(R), and L^(d) _(L). Even with thereduction of adjustable parameters due to the impedance matchingconditions or other conditions, the E-CRLH unit cell 2100 provides moreadjustable parameters than the C-CRLH or D-CRLH portion alone.

FIG. 22 plots simulation results of insertion loss of an frequencyselector configured as in FIG. 16 but with an E-CRLH TL instead of theC-CRLH TL 1604. The E-CRLH equivalent parameters are manipulated toprovide four notches in the insertion loss at 910 MHz, 2.06 GHz, 3.69GHz and 6.23 GHz, as indicated by the points m5, m6, m7 and m8,respectively, so as to block the signals in these four bands. This typeof frequency selectors based on E-CRLH unit cells may thus be termed amulti-notch filter.

By combining a frequency selector (single-notch or multiple-notch) and ahigh-pass filter, a low-pass filter, a band-pass filter, a band-stopfilter or other different filters, complex filtering functions, whichare not achievable by using conventional filters, may be provided. Basedon the flexibility in phase selection inherent in CRLH structures, thecombined filter may be tailored to provide desirable blocking andpassing of frequency bands depending on underlying applications. Theexample frequency selectors described in this document are two-portdevices and thus suited for applications in which signals with differentfrequencies are received sequentially in different time intervals sothat the signal blocking and passing occur in different time intervals.In a system in which signals with different frequencies are received atthe same time, it may be effective to have multiple branches for themultiple frequency selection. A diplexer is an example having two suchbranches, and thus is a three-port device. Combining an frequencyselector in one branch and a high-pass filter (HPF) in the other branch,for example, may provide a diplexer capability with better performancethan a conventional diplexer. Furthermore, a multiple number offrequency selectors with multiple branches may be configured for passingand blocking a multiple number of frequencies, thereby providingmultiplexer functionality.

FIG. 23 illustrates an example of a multiband frequency selector 2300,in which the input and output frequencies are represented by f₁, f₂, . .. , f_(n), the series elements are denoted by X₁, X₂, . . . , X_(n), andthe shunt elements are denoted by Y₁, Y₂, . . . , Y_(n), where n is apositive integer greater than 1. Thus, the multiband frequency selector2300 is a (n+1) port bi-directional device in that signals are allowedto transmit in either direction from the one port to the n ports or fromthe n ports to the one port. The series element X_(m) and the shuntelement Y_(m), where 1<m<n, are coupled to the same m^(th) branch toselect the frequency f_(m) for passing through the branch to an output.Each branch includes a frequency selector such as the frequencyselectors described with reference to FIGS. 8-20. Thus, the pair of theseries element X_(m) and the shunt element Y_(m) in the m^(th) branchmay both be CRLH TLs or may be a combination of an RH TL and a CRLH TL,as in the frequency selectors described with reference to FIGS. 8-20. Inaddition, a conventional filter such as a high-pass filter, a low-passfilter, a band-pass filter, a band-stop filter or another type of filtermay be used for one or more branches. Furthermore, a multi-notch filterbased on an E-CRLH structure, as illustrated in FIGS. 21 and 22, may beused for one or more branches as a frequency selector to support complexfiltering functions. Thus, the multiband frequency selector 2300 of FIG.23 is essentially a network of frequency selectors distributed on themultifurcated branches, where the frequency selectors may includeconventional filters and frequency selectors. Such a frequency selectingnetwork has (n+1) ports, where n represents the number of the frequencybands supported in the multiband operation, and is a bi-directionaldevice in that signals are allowed to transmit in either direction fromthe one port to the n ports or from the n ports to the one port.

FIG. 24 illustrates another example of a multiband frequency selector2400, in which two shunt elements are coupled to each branch, asindicated by Y₁, Y₂, . . . , Y_(n), and Y_(1a), Y_(2a), . . . , Y_(na).For example, the first shunt element may be designed to remove a signalwith a first frequency and the second shunt element in the same branchmay be designed to remove a signal with a second frequency so that thebranch passes a signal with a third frequency when the signals with thethree different frequencies are inputted to the branch. In anotherexample, the first shunt element may be designed to remove a signal witha first frequency and the second shunt element is designed to removeharmonics from a signal with a second frequency so that the branchpasses the signal with the second frequency without harmonics when thesignals with the different frequencies are inputted to the branch. Toextend the application, three or more shunt elements may be coupled toone branch, or the number of shunt elements per branch may vary frombranch to branch. Similarly, the number of series elements per branchmay be more than one and may vary from branch to branch. In thefrequency selector designs described in this document, each of theseries and shunt elements can be chosen to have an RH structure or aCRLH structure, which can be implemented using lumped components,distributed components, or a combination of both.

FIG. 25 illustrates a layout of an implementation example of a diplexer2500. This implementation is devised to provide a diplexer functionbased on the combination of the frequency selector 2504 and the HPF2508. Similar to the double-notch frequency selector 1700 of FIG. 17,the frequency selector 2504 in this example is designed to pass the 2.4GHz band and filter out the 1.9 GHz and 5 GHz bands. Minor designmodifications are made for optimization in comparison to the layout ofFIG. 17. For example, the CRLH unit cell 2502 in this case has one C_(L)capacitor, instead of two 2C_(L) capacitors of FIG. 17 which provide asymmetric unit cell, and the L-shaped RH TL has dimensions for obtainingan optimized L_(L) value different from the RH TL (L_(L)) 1712 of FIG.17. The diplexer 2500 of FIG. 25 is a three-port device similar to aconventional diplexer. An antenna or any other device may be connectedto the port P1. The port P2 of the diplexer 2500 is coupled to thefrequency selector 2504. The port P3 is coupled to the HPF 2508 that isdesigned with three capacitors C1, C2 and C3 and three microstrips 2510to filter out the 2.4 GHz band and pass the 5 GHz band. In one examplethe diplexer 2500 may have an overall footprint of 10 mm×10 mm. Thethree ports are matched to 50Ω in the example design. Note, however,that the diplexer 2500 in some implementation examples may have thecapability of matching different ports to different impedances becauseof the flexible CRLH phase response and hence the design flexibility. Asillustrated in FIG. 25, this implementation example has the HPF 2508with three microstrips 2510. The longest L-shaped microstrip is placedat the input of the HPF 2508 to rotate the phase associated with thefrequency filtered by the HPF, i.e., 2.4 GHz in this example. Thus, theHPF 2508 has an open circuit at the port P1 preventing the signal around2.4 GHz from going into the branch P1-P3.

FIG. 26 plots measurement results of the insertion loss P2-P1, theinsertion loss P3-P1 and the isolation P3-P2 for the implementationexample diplexer 2500 of FIG. 25. Around 5 GHz, the insertion loss P3-P1is close to zero and the insertion loss P2-P1 is close to −20 dB,indicating that the signal in the 5 GHz band is selectively transmittedto the port P3 through the HPF 2508 and rejected by the frequencyselector 2504. Similarly around 2.4 GHz, the insertion loss P2-P1 isclose to zero and the insertion loss P3-P1 is close to −35 dB,indicating that the signal in the 2.4 GHz band is selectivelytransmitted to the port P2 through the frequency selector 2504 andrejected by the HPF 2508. At around 1.9 GHz, both the insertion lossesP2-P1 and P3-P1 are very low, less than −25 dBm, indicating that thesignal in the 1.9 band is rejected by both the frequency selector 2504and the HPF 2508.

The examples of frequency selectors described so far may be consideredto be passive since the frequency selection is determined based on RHand/or CRLH TL designs providing proper phases without being controlledby an active component such as a diode or a switch. In differentexamples, a combination of passive frequency selectors, a combination ofRH and CRLH TLs and a combination of CRLH TLs may be structured to beactively controlled for selecting a single frequency or multiplefrequencies. These frequency selectors are thus termed active frequencyselectors in this document.

FIGS. 27A-27C illustrate an example of an active frequency selector 2700using two passive frequency selectors 2704 and 2708 connected inparallel. The active frequency selector 2700 has one input port P1 andone output port P2, and a dual signal with frequencies with f₁ and f₂may be fed simultaneously or sequentially in different time intervals tothe input port P1. These passive frequency selectors may be CRLH-based,conventional, or a combination of both. An active frequency selector canbe used for the simultaneous signal input since the frequency selectionmay be controlled by active components when two different frequencies f₁and f₂ are fed at the same time to the input port P1. In the presentexample, the frequency selector (f₁ not f₂) 2704 is configured to selectthe signal with frequency f₁ and block the signal with frequency f₂,whereas the frequency selector (f₂ not f₁) 2708 is configured to selectthe signal with frequency f₂ and block the signal with frequency f₁. TheON/OFF of the frequency selectors 2704 and 2708 may be controlled by anexternal control circuit, for example. In FIG. 27A, the frequencyselector (f₁ not f₂) 2704 is ON while the other frequency selector (f₂not f₁) 2708 is OFF. While the active frequency selector 2700 is in thisstate, only the signal with f₁ is allowed to pass through this system toreach the port P2. In FIG. 27B, the frequency selector (f₁ not f₂) 2704is OFF while the other frequency selector (f₂ not f₁) 2708 is ON. Whilethe active frequency selector 2700 is in this state, only the signalwith f₂ is allowed to pass through this system to reach the port P2. InFIG. 27C, both the frequency selector (f₁ not f₂) 2704 and the otherfrequency selector (f₂ not f₁) 2708 are ON. While the active frequencyselector 2700 is in this state, both the signals with f₁ and f₂ areallowed to pass through this system to reach the port P2.

FIG. 28 illustrates an example of an active frequency selector 2800having two CRLH TLs 2804 and 2808 connected in shunt. In this example,two different frequencies f₁ and f₂ may be fed to the port P1simultaneously or sequentially in different time intervals. These CRLHTLs 2804 and 2808 are coupled to the signal path P1-P2, each in an openshunt configuration in this example via a diode 1 2816 and a diode 22818, respectively. If shunt lines are used for connection in place ofthe diode 1 2816 and the diode 2 2818, the CRLH TL (f₁ not f₂) 2804 isconfigured to pass the signal with frequency f₁ through the path P1-P2and filter out the signal with frequency f₂, whereas the CRLH TL (f₂ notf₁) 2808 is configured to pass the signal with frequency f₂ through thepath P1-P2 and filter out the signal with frequency f₁. These CRLH TLs2804 and 2808 may be designed based on a CRLH TL design as illustratedin FIG. 8. The ON/OFF of the diode 1 2816 and the diode 2 2818 may becontrolled by an external control circuit, for example. When the diode 12816 is ON and the diode 2 2818 is OFF, the CRLH TL (f₁ not f₂) 2804 iscoupled to the path P1-P2 so that the signal with f₁ is allowed to passthrough the path P1-P2, but the signal with f₂ is blocked. When thediode 1 2816 is OFF and the diode 2 2818 is ON, the CRLH TL (f₂ not f₁)2808 is coupled to the path P1-P2 so that the signal with f₂ is allowedto pass through the path P1-P2, but the signal with f₁ is blocked. Aseries TL 2822 is included in this example along the path P1-P2. This TLmay be designed using an RH structure such as a conventional microstripor a CRLH structure so that these phases φ₃ and φ₄ have values suitedfor underlying applications to adjust the signal transmission andreflection. This TL may also be used to integrate another circuitfunction, an impedance matching network, for example.

FIG. 29 illustrates another example of an active frequency selector2900. In this example, the CRLH TLs 2904 and 2908 are coupled to thepath P1-P2 in an open shunt configuration via a diode 1 2916 and a diode2 2918, respectively, and used as an active harmonic trap foreffectively removing harmonics. The design for the shunt CRLH TLs 2904and 2908 may be similar to the shunt CRLH TL 2004 design in FIG. 20. Aseries TL 2920 is included in this example along the path P1-P2. This TL2920 may be designed using an RH structure such as a conventionalmicrostrip or a CRLH structure to have flexibility in phase andfrequency selection according to underlying applications. If a shuntline is used for connection in place of the diode 1 2916, the CRLH TL(f₁ not [2f₁, 3f₁]) 2904 is configured to pass the signal with frequencyf₁ through the path P1-P2 and filter out the harmonics 2f₁ and 3f₁. Thismay be accomplished by choosing φ₁₁=180° (open) at f₁, φ₁₂=90° at 2f₁(short) and φ₁₂=−90° (short) at 3f₁, for example. Similarly, If a shuntline is used for connection in place of the diode 2 2918, the CRLH TL(f₂ not [2f₂, 3f₂]) is configured to pass the signal with frequency f₂through the path P1-P2 and filter out the harmonics 2f₂ and 3f₂. Thismay be accomplished by choosing φ₂₁=180° (open) at f₂, φ₂₂=90° at 2f₂(short) and φ₂₂=−90° (short) at 3f₂, for example. The ON/OFF of thediode 1 2916 and the diode 2 2918 may be controlled by an externalcontrol circuit (not shown) to select the signal with f₁ or the signalwith f₂ for transmission through the path P1-P2. The TL 2920 may bedesigned with an RH TL or a CRLH TL to have adequate phases forunderlying applications. For example, a class F PA application includesthe case in which the port P1 needs to be shorted for the secondharmonic and to be open for the third harmonic. In this case, by using aCRLH TL, the phases of the TL 2920 may be chosen to be φ′₁₁=arbitrary atf₁, φ′₁₂=180° at 2f₁, φ′₂₃=90° at 3f₁, φ′₂₁=arbitrary at f₂, φ′₂₂=−180°at 2f₂, and φ′₂₃=−90° at 3f₂, for example. Similarly, an inverse class FPA application includes the case in which the port P1 needs to be openfor the second harmonic and to be shorted for the third harmonic. Inthis case, by using a CRLH TL, the phases of the TL 2920 may be chosento be φ′₁₁=arbitrary at f₁, φ′₁₂=90° at 2f₁, φ′₁₃=180° at 3f₁,φ′₂₁=arbitrary at f₂, φ′₂₂=−90° at 2f₂, and φ′₂₃=−180° at 3f₂, forexample.

As mentioned earlier, obtaining high linearity and/or high efficiency isone of the main goals of PA designs. When used in a conventionalsingle-band PA, an output matching network (OMN) is generally designedto match the output impedance of a transistor to the output load so asto maximize the output power transfer from the transistor to the load.Thus, the OMN may be tailored to improve linearity and efficiency.Similarly, an input matching network (IMN) may be designed to match theinput impedance of the transistor to the input load. Thus, in aconventional single-band PA, an IMN and an OMN may be utilized foroptimizing efficiency and linearity of the PA over the single band.However, for multiband operations other circuit elements may need to beadded, designed or optimized to meet efficiency, linearity and otherconsiderations over multiple frequency bands.

FIG. 30 illustrates a block diagram of a first power amplificationsystem 3000 for multiple frequency bands. This power amplificationsystem 3000 represents an example of a multiband PA. The first system3000 has multiple input ports and multiple output ports for handlingmultiple frequency bands individually, and includes a multiband biascircuit 3004, a frequency selecting module 1 3008 on the input side,another frequency selecting module 2 3012 on the output side, an IMN3016 including IMN (f₁), IMN (f₂), . . . , IMN (f_(n)) for inputimpedance matching for respective frequency bands, an OMN 3020 includingOMN (f₁), OMN (f₂), . . . , OMN (f_(n)) for output impedance matchingfor respective frequency bands, and a transistor 3024 that receives biassignals from the multiband bias circuit 3004 and is coupled to thefrequency selecting module 1 3008 and the frequency selecting module 23012. The transistor 3024 is a multiband transistor operable for themultiple frequency bands and may include multiple basic transistors forhandling the frequency range and power levels considered for underlyingmultiband applications. The multiband bias circuit 3004 may includemultiple basic bias circuits for biasing different terminals of thetransistor 3004. Typically, a drain bias and a gate bias are needed fora Metal-Oxide-Semiconductor Field Effect Transistor (MOSFET) type oftransistor; a collector bias and a base bias are needed for a BipolarJunction Transistor (BJT) type of transistor or Heterojunction BipolarTransistor (HBT). In general, a bias circuit is designed to deliver DCpower to the transistor while being transparent to the RF signal so asto prevent the RF signal leakage into the bias circuit that may degradethe efficiency and linearity of the PA in which the bias circuit isincluded. To perform the similar operation for two or more frequencybands, the multiband bias circuit 3004 may include a multiband biascircuit as illustrated in FIG. 6.

When different signals with different frequencies f₁, f₂, . . . , andf_(n) are inputted simultaneously, the frequency selecting module 1 3008is configured to select the signal with one frequency, f₁, f₂, . . . ,or f_(n), from the input signals coming out of the IMNs 3016. One roleof this frequency selecting module 1 3008 on the input side is to send asignal with one frequency in a given time interval to the transistor3024, instead of mixed signals with different frequencies. Generally, ifsignals with two different frequencies are sent at the same time to atransistor, several intermodulation products may be outputted from thetransistor. Such an intermodulation arises from non-linearity betweenthe input and the output of a transistor, resulting in possibledetrimental effects. The intermodulation associated with the transistor3024 may thus be avoided by use of the frequency selecting module 1 3008for the case in which multiple signals with different frequencies areinputted simultaneously. Further, the frequency selecting module 1 3008is configured to prevent signal reflections into wrong paths so as tominimize the signal mixing and maximize the power transfer, therebyimproving the linearity and efficiency. When multiple signals withdifferent frequencies f₁, f₂, . . . , and f_(n) are inputtedsequentially in different time intervals, the intermodulation of thetransistor 3024 does not occur under normal conditions. However, thefrequency selecting module 1 3008 is used to prevent signal reflectionsinto wrong paths so as to minimize the signal mixing and maximize thepower transfer, thereby improving the linearity and efficiency. Thefrequency selecting module 2 3012 on the output side is configured tosend a signal with a certain frequency f_(m) (1≦m≦n) to the m^(th)output port, and/or remove harmonics from the output signals from thetransistor 3024.

FIG. 31 illustrates a block diagram of a second power amplificationsystem 3100 for multiple frequency bands. The second system 3100 has asingle input port and a single output port for handling multiplefrequency bands in one channel, and includes a multiband bias circuit3104, a frequency selecting module 1 3108 on the input side for sendingthe signal with one frequency f_(m) to the m^(th) branch (1≦m≦n),another frequency selecting module 2 3112 on the output side to select asignal with one frequency and send the selected signal to the outputport, an IMN 3116 including IMN (f₁), IMN (f₂), . . . , IMN (f_(n)) forinput impedance matching for respective frequency bands, an OMN 3120including OMN (f₁), OMN (f₂), . . . , OMN (f_(n)) for output impedancematching for respective frequency bands, and multiple transistors 3124,each receiving bias signals from the multiband bias circuit 3104 andcoupled to one of the IMN (f₁), IMN (f₂), . . . , IMN (f_(n)) and to oneof the OMN (f₁), OMN (f₂), . . . , OMN (f_(n)). Each of the transistors3124 is a single-band transistor and may include multiple basictransistors for handling the frequency range and power levels consideredfor underlying applications. These transistors 3124 may be implementedusing separate discrete packages or banks of segmented basic transistorson an IC chip. The multiband bias circuit 3104 may include multiplebasic bias circuits for biasing multiple transistors 3104 and differentterminals of each of the transistors 3104. The multiband bias circuit3104 may include a multiband bias circuit such as shown in FIG. 6.

In the second system 3100 of FIG. 31, multiple signals with differentfrequencies f₁, f₂, . . . , and f_(n) may be inputted simultaneously orsequentially in different time intervals. The frequency selecting module1 3108 on the input side in the second system 3100 of FIG. 31 works inthe similar way as the frequency selecting module 2 3012 on the outputside in the first system 3000 of FIG. 30; and the frequency selectingmodule 2 3112 on the output side in the second system 3100 of FIG. 31works in the similar way as the frequency selecting module 1 3008 on theinput side in the first system 3000 of FIG. 30.

The second power amplifying system 3100 of FIG. 31 represents an exampleof a multiband PA. In addition, by using an array of single-band PAs,each accommodating the IMN (f_(m)), one of the transistors 3124 handlingthe f_(m) band and the OMN (f_(m)), where 1≦m≦n, a power amplifyingsystem in a compact design may be realized for multiband operations.

The second system 3100 of FIG. 31 may be modified for the case of oneinput port and multiple output ports by removing the frequency selectingmodule 2 3112 from the output side. Similarly, this system 3100 may bemodified for the case of multiple input ports and one output port byremoving the frequency selecting module 1 3108 from the input side.

The frequency selecting module in the above power amplification systems3000 and 3100 of FIGS. 30 and 31 may be configured to include amultiband frequency selector such as illustrated in FIG. 23 or FIG. 24.Further, the frequency selecting module may be configured to include anactive frequency selector such as illustrated in FIG. 28 or FIG. 29.Furthermore, a combination of a multiband frequency selector and anactive frequency selector may be utilized in one frequency selectingmodule.

FIG. 32 illustrates an example of a dual-band PA 3200 based on the firstpower amplification system 3000 of FIG. 30 using active and passivefrequency selectors for the first and second frequency modules. Thisdual-band PA 3200 has two input ports P1 and P2 and two output ports P3and P4. The dual signals with frequencies f₁ and f₂ may be fedsimultaneously or sequentially in different time intervals to the inputports P1 and P2, respectively. In this example, an active frequencyselector 3204, such as the active frequency selector 2800 of FIG. 28, isplaced on the input side; and a harmonic trap 3212, such as the activefrequency selector 2900 of FIG. 29, is placed on the output side. TheON/OFF of the diodes in the active frequency selector 3204 is controlledto select the signal with either f₁ or f₂, which is to be sent to atransistor 3208 for amplification. The transistor may be a MetalSemiconductor Field Effect Transistor (MESFET), a Pseudomorphic HighElectron Mobility Transistor (pHEMT), a Heterojunction BipolarTransistor (HBT) or of other suitable technologies. The harmonic trap3212 is designed to remove the second and third harmonics for f₁ and f₂in this example. Specifically, by controlling the ON/OFF of the diodesin the harmonic trap 3212, the harmonics 2f₁ and 3f₁ may be removed whenthe signal with f₁ is outputted from the transistor 3208 or theharmonics 2f₂ and 3f₂ may be removed when the signal with f₂ isoutputted from the transistor 3208. The output of the harmonic trap 3212is coupled to the frequency selector (f₁ not f₂) 3216 and the frequencyselector (f₂ not f₁) 3220 that are connected in a diplexer form. Thus,the signal with f₁ is selected to pass through the upper branch to theoutput port P3 and the signal with f₂ is selected to pass through thelower branch to the output port P4. Each of these frequency selectors onthe output side may include a passive frequency selector described withreference to FIGS. 8-22. The IMN 1 3224 is coupled to the input port P1,the IMN 2 3228 is coupled to the input port P2, the OMN 1 3232 iscoupled to the output port P3, and the OMN 2 3236 is coupled to theoutput port P4.

FIG. 33 illustrates another example of a dual-band PA 3300 based on thefirst power amplification system 3000 of FIG. 30. This dual-band PA 3300has two input ports P1 and P2 and two output ports P3 and P4, andincludes two passive frequency selectors 3304 and 3308 on the input sideand two passive frequency selectors 3316 and 3320 on the output side.Examples of passive frequency selectors described with reference toFIGS. 8-22 may be utilized to construct these passive frequencyselectors 3304, 3308, 3316 and 3320. The IMN 1 3324 is coupled to theinput port P1, the IMN 2 3328 is coupled to the input port P2, the OMN 13332 is coupled to the output port P3, and the OMN 2 3336 is coupled tothe output port P4. Signals with f₁ and f₂ are fed at the input ports P1and P2, respectively. Due to the use of the passive frequency selectors3304 and 3308 on the input side in this example, the signals may need tobe inputted sequentially in different time intervals to avoidintermodulation which may occur if signals from both input ports P1 andP2 are inputted simultaneously and reach the transistor simultaneously.The frequency selector (f₁ not f₂) 3304 selects the signal with f₁ butnot the signal with f₂ and blocks the signal f₂ reflection, while thefrequency selector (f₂ not f₁) 3308 selects the signal with f₂ but notthe signal with f₁ and blocks the signal f₁ reflection. The selectedsignal with f₁ or f₂ is then amplified by the transistor 3312. Theoutput frequencies are selected at the output branches having thefrequency selector (f₁ not f₂) 3316 on one branch to select f₁ and thefrequency selector (f₂ not f₁) 3320 on the other branch to select f₂.Thus, the signal with f₁ is outputted to the output port P3; and thesignal with f₂ is outputted to the port P4.

FIG. 34 illustrates another example of a dual-band PA 3400 based on thefirst power amplification system 3000 of FIG. 30. This dual-band PA 3400has two input ports P1 and P2 and two output ports P3 and P4, andincludes an active frequency selector 3404, such as the active frequencyselector 2800 of FIG. 28, on the input side and two passive frequencyselectors 3416 and 3420 on the output side. The IMN 1 3416 is coupled tothe input port P1; the IMN 2 3420 is coupled to the input port P2; theOMN 1 3432 is coupled to the output port P3; and the OMN 2 3436 iscoupled to the output port P4. The signals with f₁ and f₂ may beinputted from the input ports P1 and P2, respectively, simultaneously orsequentially in different time intervals. By controlling the ON/OFF ofthe diodes in the active frequency selector 3404, the signal withfrequency f₁ or the signal with frequency f₂ is selected. The selectedsignal with f₁ or f₂ is then amplified by the transistor 3412. Theoutput frequencies are selected at the output branches having thefrequency selector (f₁ not f₂) 3416 on one branch to select f₁ and thefrequency selector (f₂ not f₁) 3420 on the other branch to select f₂.

As mentioned earlier, it is generally preferred that PAs have highlinearity and/or efficiency in order to meet various specifications andachieve a sufficient performance level in communication systems. Insystems such as the Global System for Mobile (GSM) communications, PAsare preferred to have high efficiency, but linearity considerations maybe less stringent because constant envelope signals are used therein. Inother systems, such as the Code Division-Multiple Access (CDMA) systems,non-constant envelope signals are used, and thus linearityconsiderations are likely to be stringent. Generally, efficiency andlinearity have a trade-off relationship in a PA. FIG. 35 illustrates theI-V characteristics of a MESFET as an example. The bias points ofvarious class operations are indicated in this figure. A MESFET isdriven by a negative gate-to-source voltage Vgs having the maximumcurrent at Vgs=0. As the Vgs is lowered until the pinch-off at Vgs=Vp,the drain-to-source current Ids decreases. Below the Vgs=Vp line is thecut-off region. A class-A PA has the bias point with a high Vgs valueand hence a high Ids value. This equates to high output power and hencehigh linearity. However, the class-A operation is mostly in an ON-statein spite of varying signal envelopes, thereby resulting in lowefficiency due to prolonged power loss. Thus, a class-A PA is consideredto be a linear amplifier but generally has a poor efficiency rating. Aclass-C is operated with a very low Vgs value below the pinch-off forspecial applications. Class-B, class-E, class-F and their inverses havebias points with Vgs values close to pinch off, being capable ofproviding ON/OFF controls for varying signal envelopes so as to reducepower loss and thus increase efficiency compared to class-A. However,the linearity in these PAs may not be sufficient for certainapplications due to the low Ids current and hence low output power. Insome cases, PAs may be designed to operate in two modes: class-AB andclass-F, for example, by changing the bias point of the PA. This type ofoperation may contribute to achieving high efficiency and linearity inspite of the trade-off relationship. In applications where both moderateefficiency and linearity are preferred, a class-AB PA may be chosensince the bias point is located between the class-A and the classes-B, Eand F. A class-J PA may generally be viewed as a class-AB PA plus aharmonic trap used to remove harmonics. In a class-J scheme, it ispossible to have both the second and third harmonics shorted. Thisallows the drain (collector) current and voltage to have sinusoidalwaveforms, which will enhance linearity of the class-J PA.

The following PA architectures in this document are based on a class-JPA. By using a CRLH TL for the harmonic trap in the class-J PA, anexample of a class-J MTM PA is devised and described as below. UsingCRLH structures may contribute to a circuit size reduction and adecrease in power loss, which may consequently improve efficiency.Additionally, biasing the PA close to the class-AB bias point isexpected to lead to better linearity than class-B, E, F and theirinverses. Note, however, that the similar harmonic trapping scheme basedon CRLH structures can be used for other classes as well depending onefficiency and linearity considerations of underlying applications.

FIG. 36A-36C illustrate schematic diagrams of three differentconfigurations for removing harmonics in a PA. FIG. 36A illustrates aconventional configuration of a class-J PA using a conventional harmonictrap 3616, which includes multiple RH TLs for removing multipleharmonics individually. This PA includes an IMN 3604 on the input sideand an OMN 3612 on the output side, and the harmonic trap 3616 coupledto the transistor drain (or collector) terminal through an RH TL 3608such as a microstrip. FIG. 36B illustrates a configuration of a class-JMTM PA having an MTM harmonic trap 3620 for removing multiple harmonicsbased on one CRLH TL in place of multiple TLs in FIG. 36A. The frequencyselector 2000 of FIG. 20, for example, may be utilized for the harmonictrap 3620 to remove the second and third harmonics, thereby resulting inless components than the conventional harmonic trap 3616. Such anoverall circuit size reduction may consequently reduce the overall powerloss, thereby improving efficiency. FIG. 36C illustrates a class-J MTMPA having an OMN-integrated MTM harmonic trap 3624. In this integrationexample, the RH portion of the CRLH TL, i.e. an RH inductance L_(R) andan RH capacitance C_(R), are structured for optimizing the outputimpedance matching in addition to providing adequate phases as part ofthe CRLH TL in the MTM harmonic trap 3624.

FIG. 37 illustrates a layout of an implementation example 3700 of theclass J MTM PA with the MTM harmonic trap 3620 of FIG. 36B. A MESFET3704 is used in this example due to availability of the discrete packagefor the underlying frequency range. However, the transistor may be of adifferent type with a different fabrication technique. FIG. 37schematically illustrates additional discrete components used to realizethis PA. The gate of the MESFET 3704 is biased by the gate bias circuit3708, which includes two capacitors, a resistor and an RF choke. Thedrain of the MESFET 3704 is biased by the drain bias circuit 3712, whichincludes a capacitor and an RF choke. Although conventional single-bandbias circuits are used for this implementation example 3700, the biascircuit as illustrated in FIG. 6 may be used for one or both of the gateand drain bias circuits. The IMN 3716 includes a series RH TL and ashunt RH TL, indicated as RH TL 1 3720 and RH TL 2 3724, respectively.As explained earlier with reference to FIGS. 1F and 1G, the combinationof a series RH TL and a shunt RH TL may be modeled with an RH seriesinductance L_(R) and an RH shunt capacitor C_(R). By structuring thelengths and widths of the RH TLs 3720 and 3724, these equivalent circuitparameters may be adjusted to provide the optimum input impedancematching. A DC bock 1 3728 is included in the path along the RH TL 13720 to block DC signals. Similarly, the OMN 3732 includes a series RHTL and a shunt RH TL, indicated as RH TL 3 3736 and RH TL 4 3740,respectively. By structuring the lengths and widths of the RH TLs 3736and 3740, the equivalent circuit parameters may be adjusted to providethe optimum output impedance matching. A DC block 2 3744 is included inthe path along the RH TL 3 3736 to block DC signals. The frequencyselector 3748 plays a role of the MTM harmonic trap. Four shunt LHcapacitors, i.e., two 2C_(L)s and two C_(L)s, three shunt L_(L)inductors, i.e., three L_(L)s, and the RH TL 5 3752 are configured toprovide three CRLH unit cells in this example. These parameter valuesare manipulated to provide the harmonic trap function, i.e., to removethe second and third harmonics and pass the signal with the primaryfrequency by using the CRLH design scheme for the harmonic trapdescribed with reference to FIG. 20, in which the CRLH phase curve isfit to three points. The phases of the RH TL 6 3756 may also bedetermined using the RH design scheme described with reference to FIG.20 in that the input port P1, which is coupled to the drain terminal ofthe MESFET 3704, has an open circuit for odd harmonics and a shortcircuit for even harmonics, as generally considered for class-Japplications. The phases of the RH TL 6 3756 may be adjusted differentlydepending on underlying applications to adjust the signal transmissionand reflection. The efficiency may be improved due to the reduced sizeof the frequency selector 3748 compared to the conventional RH harmonictrap in FIG. 36A. Furthermore, sufficient linearity may also be achieveddue to the use of the class AB bias.

FIG. 38 plots measurement results of Power Added Efficiency (PAE) andoutput power (Pout) as a function of input power (Pin) of theimplementation example 3700 of the Class-J MTM PA of FIG. 37. In an RFPA, PAE is defined as the ratio of the difference between Pout and Pinto the DC power consumed, whereas efficiency is defined as the ratio ofPout to the DC power consumed. The Pout versus Pin curve in FIG. 38 isapproximately linear, indicating good linearity, in this measurementrange in which Pout=18 dBm corresponds to power saturation. Linearitymay also be evaluated by using Error Vector Magnitude (EVM), which is ameasure of how far the points are from the ideal lattice points,expressed as a percentage. Generally, an EVM diagram illustrates thatthe fixed lattice points correspond to non-distortion of the signalforms and the distortions are quantized by the deviations from thelattice points. Thus, as linearity improves, the EVM value decreases.The EVM value of 0% corresponds to non-distortion and thus to ideallinearity. The measured EVM for this example is about 5.6%, which isconsidered of moderately good linearity in this application. The PAEversus Pin curve in FIG. 38 indicates that this PA has a peak PAE ofabout 52% and the overall PAE in the linear region is greater than 30%.This PAE value is significantly high as compared to a conventionalclass-AB PAE value of about 15%.

FIG. 39 illustrates a layout of an implementation example 3900 of theclass-J MTM PA with an OMN-integrated MTM harmonic trap 3624 of FIG.36C. A pHEMT is used in this example due to the availability of thediscrete package for the underlying frequency range. However, thetransistor may be of a different type with a different fabricationtechnique. FIG. 39 also illustrates additional discrete components usedto realize this PA. The base of the pHEMT 3904 is biased by the basebias circuit 3908, which includes a capacitor and a resistor. Thecollector of the pHEMT 3904 is biased by the collector bias circuit3912, which includes a capacitor. Each of these bias circuits includes aradial quarter-wavelength RH TL 3916/3917 and a stub-formquarter-wavelength RH TL 3918/3919, providing a total of half-wavelengthfor transforming the impedance from an open to an open as illustrated inFIG. 4. Use of a radial quarter-wavelength RH TL generally contributesto increasing the bandwidth. Although these bias circuits are designedbased on the conventional single-band bias scheme, the bias circuit 600based on a CRLH structure as shown in FIG. 6 may be used for one or bothof the base and collector bias circuits. The IMN 3924 includes a seriesRH TL and a shunt RH TL, similar to the IMN 3716 in the class-J MTM PAimplementation of FIG. 37. As explained earlier with reference to FIGS.1F and 1G, the combination of a series RH TL and a shunt RH TL may bemodeled with an RH series inductance L_(R) and an RH shunt capacitorC_(R). By structuring the lengths and widths of these two RH TLs, theseequivalent circuit parameters may be adjusted to provide the optimuminput impedance matching. A DC block 1 3928 is included in the IMN 3924to block DC signals. The OMN-integrated MTM frequency selector 3932 isan implementation example of the OMN-integrated MTM frequency selector3624 in FIG. 36C, including an RH TL 1 3936 and an RH TL 2 3940. A DCbock 2 3944 is included in the path along the RH TL 1 3936 to block DCsignals. Four shunt LH capacitors, i.e., two 2C_(L)s and two C_(L)s,three shunt L_(L) inductors, i.e., three L_(L)s, and the RH TL 2 3940are configured to provide three CRLH unit cells in this example. Thesecapacitance and inductance values, as well as the dimensions of the RHTL 2 3940, are adjusted so that the equivalent circuit parameters forthe RH portion, C_(R) and L_(R), provide the optimum output impedancematching as an OMN and at the same time the equivalent circuitparameters for the LH and RH portions, C_(L), L_(L), C_(R) and L_(R),provide the harmonic trap function, i.e., to filter out the second andthird harmonics and pass the signal with the primary frequency. A CRLHdesign scheme for a harmonic trap may be similar to the one describedwith reference to FIG. 20, in which the CRLH phase curve is fit to threepoints. The phases of the RH TL 1 3936 may also be determined using theRH design scheme described with reference to FIG. 20 in that the inputport P1, which is coupled to the collector terminal of the pHEMT 3904,is open for odd harmonics and shorted for even harmonics, as generallyconsidered for class-J applications.

Alternatively, the phases can be adjusted so that the collector terminalof the transistor is shorted for odd harmonics and open for evenharmonics, shorted for all harmonics, or open for all harmonics,depending on underlying applications.

The design for this class-J MTM PA implementation 3900 of FIG. 39involves integration of an OMN into an MTM harmonic trap as illustratedin FIG. 36C. As previously mentioned, the use of an MTM harmonic trapmay reduce the overall circuit size: in contrast to the multiple RH TLsused in a conventional design for a harmonic trap, such as in FIG. 36A,one CRLH stub may be designed to filter out the second and thirdharmonics as in FIGS. 36B and 36C. In addition, the implementationexample 3900 of FIG. 39 integrates an OMN into a CRLH TL, which is partof the MTM harmonic trap, further reducing the circuit size andcomponent count. A reduction in overall circuit size and complexitygenerally reduces power loss, thereby leading to improved efficiency.

FIG. 40 plots measurement results of Pout versus Pin for theimplementation example 3900 of the Class-J MTM PA of FIG. 39. A basebias of Vbe=1V and a collector bias of Vce=3.5V are chosen for thisclass-AB case. The plot indicates that good linearity is obtained withthe Pout up to about 15 dBm in this application. Note that the presentimplementation of the MTM class-J PA has some gain even in the low Pinregion.

FIG. 41 plots measurement results of PAE versus Pout for theimplementation example 3900 of the class-J MTM PA of FIG. 39. The plotindicates that the measured peak PAE is about 60% at about Pout=20 dBm.The measured EVM values are added in this figure, indicating that an EVMvalue of less than 3% is obtained up to a PAE value of about 30% with aPout value of about 14 dBm. Note that the EVM values of 3% or less aregenerally considered for communication systems for WiFi applications, inwhich the Pout is backed-off to this 3% EVM point to regain linearitybut with reduced efficiency.

FIG. 42 plots measurement results of PAE versus Pin for theimplementation example 3900 of the class-J MTM PA of FIG. 39. The plotindicates that the measured PAE is about 30% with an EVM value of about3% at a Pin value of about −2 dBm. The measured peak efficiency is about60% at a Pin value of about 7 dBm.

FIG. 43 plots simulation results of PAE versus frequency for amonolithic microwave integrated circuit (MMIC) implementation of theclass-J MTM PA 3900 of FIG. 39 with a driver PA coupled. OFDM signalsare used in the simulation and the EVM for this case is about 3%. Theplots indicate that good efficiency is obtained over a wideband, atleast from 1.7 GHz to 2.7 GHZ, using the present class-J MTM PA design.

Phase distortion may occur at the output of the transistor due to theinherent non-linearity. Additional circuits or components may beincorporated in a PA to minimize the phase distortion. FIG. 44illustrates a schematic diagram of a configuration example using a CRLHTL 4404 with a varactor therein. The CRLH TL 4404 is located on theinput side of the transistor 4408 in this example. The magnified view ofthe CRLH TL 4404 is shown in the dashed box, illustrating the use of aCRLH unit cell such as in FIG. 1B. This CRLH TL 4404 includes two RH/2TLs as an example to provide the RH parameters C_(R) and L_(R), butlumped elements may be used instead. A different type of a CRLH unitcell may also be used with lumped elements, distributed elements or acombination of both. The CRLH TL 4404 includes the varactor 4420 insteadof a capacitor to provide the adjustable C_(L). After the OMN 4412, theoutput power and/or associated signals are detected by a detector 4416in this example. A feedback loop is provided from the detector 4416 tocontrol the varactor 4420 inside the CRLH TL 4404. According to thedetected power and/or associated signals by the detector 4416, thecapacitance value of the varactor 4420 is varied so that the phase ofthe CRLH TL 4404 and hence the input phase is adjusted. The other CRLHparameters, such as L_(R), C_(R) and L_(L), may be kept constant orvaried for optimization. Unlike the RH phase response, which is negativeand linear in frequency, the CRLH phase response is positive or negativeand non-linear in frequency. Therefore, arbitrary phase distortion maybe adjusted using the CRLH TL 4404 with the varactor 4420 at the inputside of a PA.

In another example, an IMN may be integrated in a CRLH TL, which therebyperforms both the input impedance matching and the phase distortionminimization. The integration may be realized in the similar way as forthe OMN-integrated MTM frequency selector described with reference toFIGS. 36C and 39.

Note that the MTM PA examples described above may also be optimized forWLAN applications such as 802.11b, g with 64-QAM OFDM. Although the twoclass-J MTM PA implementations presented herein are designed tooptimally operate at one frequency, e.g., 2.4 GHz, it is possible todesign them at a different selected frequency and, therefore, may makethese MTM PAs suitable for applications such as WCDMA, LIE, WLAN802.11x, GSM, or other applications.

While this document contains many specifics, these should not beconstrued as limitations on the scope of an invention or of what may beclaimed, but rather as descriptions of features specific to particularembodiments of the invention. Certain features that are described inthis document in the context of separate embodiments can also beimplemented in combination in a single embodiment. Conversely, variousfeatures that are described in the context of a single embodiment canalso be implemented in multiple embodiments separately or in anysuitable subcombination. Moreover, although features may be describedabove as acting in certain combinations and even initially claimed assuch, one or more features from a claimed combination can in some casesbe excised from the combination, and the claimed combination may bedirected to a subcombination or a variation of a subcombination.

Only a few implementations are disclosed. Variations and enhancements ofthe disclosed implementations and other implementations can be madebased on what is disclosed and illustrated.

What is claimed is:
 1. A power amplifying system configured to operatefor a plurality of frequency bands, comprising: one or more transistorsconfigured to receive input signals in the plurality of frequency bandsand configured to amplify the input signals to generate output signalsin the plurality of frequency bands; an output matching network, locateddownstream from the one or more transistors along a data pathway,configured to perform output impedance matching for the output signalsin the plurality of frequency bands; and a frequency selecting module,located downstream from the output matching network along the datapathway, that is coupled to the output matching network and configuredto direct the output signal in each of the plurality of frequency bandsto an output signal path associated with the frequency band; wherein thefrequency selecting module includes at least one filtering elementconfigured to direct signals by filtering signals based on theirfrequency, wherein certain signals are allowed to pass and other signalsare removed via filtration.
 2. The system of claim 1, furthercomprising: a plurality of output ports that are coupled to the outputmatching network and configured to output the output signals in theplurality of frequency bands, respectively; wherein the frequencyselecting module is configured to direct the output signal in each ofthe plurality of frequency bands to the output signal path associatedwith the frequency band and coupled through the output matching networkto one of the plurality of output ports that outputs the output signalin the frequency band.
 3. The system of claim 1, further comprising: anoutput port that is coupled to the frequency selecting module andconfigured to output the output signals in the plurality of frequencybands; wherein the one or more transistors comprise a plurality ofsingle-band transistors that are configured to operate for the pluralityof frequency bands, respectively; wherein the output matching network isconfigured to receive the output signals from the plurality ofsingle-band transistors, and configured to process and send the outputsignals to the frequency selecting module; and wherein the frequencyselecting module is configured to direct the output signal in each ofthe plurality of frequency bands to the output signal path associatedwith the plurality of frequency bands and coupled to the output port. 4.The system of claim 1, wherein the frequency selecting module includes aplurality of frequency selectors, each of which is configured totransmit a signal in a frequency band associated with the frequencyselector, and remove signals in one or more frequency bands that aredifferent from the frequency band associated with the frequencyselector.
 5. A power amplifying system configured to operate for aplurality of frequency bands, comprising: an input matching networkconfigured to perform input impedance matching for input signals in theplurality of frequency bands; a frequency selecting module that iscoupled to the input matching network and configured to direct the inputsignal in each of the plurality of frequency bands to an input signalpath associated with the corresponding frequency band; and one or moretransistors configured to receive the input signals in the plurality offrequency bands and configured to amplify the input signals to generateoutput signals in the plurality of frequency bands; wherein thefrequency selecting modules includes at least one filtering elementconfigured to direct signals by filtering signals based on theirfrequency, wherein certain signals are allowed to pass and other signalsare removed via filtration wherein, along a data pathway, the frequencyselection module is upstream from the input matching network, and theone or more transistors is located downstream from the input matchingnetwork.
 6. The system of claim 5, further comprising: a plurality ofinput ports that are coupled to the input matching network andconfigured to receive the input signals in the plurality of frequencybands, respectively; wherein the input matching network is configured toreceive the input signals from the plurality of input ports andconfigured to process and send the input signals to the frequencyselecting module; and wherein the frequency selecting module isconfigured to direct the input signal in each of the plurality offrequency bands to the input signal path associated with the pluralityof frequency bands and coupled to the at least one transistor.
 7. Thesystem of claim 5, further comprising: an input port that is coupled tothe frequency selecting module and is configured to receive the inputsignals in the plurality of frequency bands; wherein the one or moretransistors comprise a plurality of single-band transistors that areconfigured to operate for the plurality of frequency bands,respectively; wherein the frequency selecting module is configured toreceive the input signals in the plurality of frequency bands from theinput port and is configured to direct the input signal in each of theplurality of frequency bands to the input signal path associated withthe frequency band and coupled through the input matching network to oneof the plurality of single-band transistors that operates for thefrequency band.
 8. The system of claim 5, wherein the frequencyselecting module include a plurality of frequency selectors, each ofwhich is configured to transmit a signal in a frequency band associatedwith the frequency selector, and remove signals in one or more frequencybands that are different from the frequency band associated with thefrequency selector.
 9. A method of making a power amplifying systemoperable for a plurality of frequency bands, the method comprising:receiving input signals in the plurality of frequency bands; amplifyingthe input signals to generate output signals in the plurality offrequency bands; matching an output impedance for the output signals inthe plurality of frequency bands; directing, after the matching, theoutput signal in each of the plurality of frequency bands to an outputsignal path associated with the frequency band; wherein said directingcomprises filtering the output signal, wherein certain signals areallowed to pass and other signals are removed via filtration.
 10. Thesystem of claim 9, wherein the system includes a plurality of outputports that are configured to output the output signals in the pluralityof frequency bands, respectively, and said directing further comprisesdirecting the output signal to one of the plurality of output ports thatoutputs the output signal in the frequency band.
 11. A method of makinga power amplifying system operable for a plurality of frequency bands,the method comprising: receiving input signals in the plurality offrequency bands; directing the received input signals in each of theplurality of frequency bands to an input signal path associated with theplurality of frequency bands; matching, after the directing, an inputimpedance of the received and directed input signals; and amplifying thedirected input signals to generate output signals in the plurality offrequency bands; wherein said directing comprises filtering the receivedinput signals, wherein certain signals are allowed to pass and othersignals are removed via filtration.